Tag tracking

ABSTRACT

The present invention provides a tracking system in which a mobile tag having an unknown position, which tag is to be tracked in space over time, transmits a signal comprising a pair of tones at different frequencies. The transmitted signal is received at each of three receivers, each having a known location, where the phase of each of the tones within the signal is measured. The measured phases are passed to a processing unit which determines the position of the tag at the time of transmission of the signal on the basis of the difference between the measured phases of the two tones. The tracking system operates over a defined finite range to track the position of the mobile tag uniquely in space.

[0001] This invention relates to a method and apparatus for trackingmoving objects. The invention has particular but not exclusive relevanceto the tracking of competitors of a race using electronic tags which arecarried by the competitors and which transmit signals that are detectedby a tracking system.

[0002] There is a requirement for a system to track the movement ofcompetitors in a race or similar sporting event to provide movement datafor use in race reconstruction and simulation services. Sporting eventsof interest include for example, horse racing, dog racing, motor racing,golf etc. Such a tracking system requires any device to be carried on orby the competitors in the racing event to be as small and light andunobtrusive as possible so as not to impede the competitors in the raceor event. Also, in order to reduce costs and operating difficulties, anyRF signals used by the system would preferably be within a frequencyband in which no licence is required and in a band in which transmissionis permitted. Further, the position accuracy should be sufficient thatthe data generated accurately describes the position of the competitorsrelative to one another. The tracking range of the system must also beable to cover the size of the venue at which the racing event is to takeplace.

[0003] U.S. Pat. No. 5,045,861 describes a mobile receiver which ismounted in, for example, a motor vehicle, and which is operable toreceive signals transmitted from a number of fixed transmitter stations.These received signals are then transmitted to a fixed receiver whichalso receives the signals from the transmitter stations. The signalstransmitted by the transmitter stations are single tone signals and thefixed receiver calculates the position of the mobile receiver from thedifference in phase between the signals received from the mobilereceiver and the signals received directly from the fixed transmitterstations.

[0004] The system described in US '861 has a number of practicalproblems which make it unsuitable for use in a system for tracking themovement of competitors in a race or similar sporting event. One of themain problems is that when a single tone is transmitted between thetransmitter and the mobile receiver, the distance between the two mustbe less than the wavelength of the transmitted tone if an absoluteposition measurement is to be determined. If this is not the case, thena phase ambiguity problem arises. In an application such as horse racingor dog racing, the measurement range may need to be between a fewhundred metres and a few kilometres. This requires a transmissionfrequency in the kilohertz or megahertz part of the radio spectrum.However, use of this part of the radio spectrum is highly regulatedmaking it impractical to use these frequencies. One solution to thisproblem is to use a higher frequency and to track the position of themobile receiver as it moves from one wavelength of the transmitted toneto the next. However, this requires the absolute position of the mobilereceiver to be known at some initial starting point. Another alternativeis to lower the frequency of the transmitted tones, however this reducesthe resolution of the position measurement making it difficult todistinguish between the different competitors of the race.

[0005] The present invention aims to provide an alternative system fortracking objects using phase measurements which at least alleviates oneor more of these problems.

[0006] According to one aspect, the present invention provides aposition determining system comprising:

[0007] a tag and a plurality of base stations, wherein the tag and theplurality of base stations are arranged so that upon the transmission ofa signal comprising first and second frequency components having afrequency spacing therebetween by one of them, there is generated aplurality of received signals each associated with a respectivetransmission path between a respective base station and the tag;

[0008] means for processing each received signal to determine a phasemeasurement for the first frequency component and a phase measurementfor the second frequency component;

[0009] means for calculating a phase difference measurement for eachreceived signal from the corresponding determined phase measurements;and

[0010] means for determining the relative position of the tag and thebase stations on the basis of the calculated phase differencemeasurements.

[0011] In a preferred embodiment, the tag is a transmit-only devicewhich is operable to transmit the signal having the first and secondfrequency components, since this simplifies the design or the tag.

[0012] In another preferred embodiment, separate fixed tags are providedwhich operate in the same way as the or each mobile tag and are used tore-reference the signals received by the base stations to a commonreference clock signal. In this way, the receivers do not need to besynchronised with each other. Preferably the or each fixed tag islocated at the same location as a corresponding one of the basestations, since this reduces the computational complexity of theposition calculations.

[0013] In another preferred embodiment, the transmitted signal comprisesat least three frequency components in which the spacing between thefirst and second frequency components is greater than the spacingbetween the second and third frequency components, whereby a coarseposition measurement can be obtained using the phase differencemeasurements from the first and second frequency components and a fineposition measurement can be obtained from the phase differencemeasurements obtained from the second and third frequency components.

[0014] Various other advantageous features and aspects of the presentinvention will become apparent from the following detailed descriptionof exemplary embodiments which are described with reference to theaccompanying drawings in which:

[0015]FIG. 1 is a schematic drawing showing a tracking system of a firstembodiment for tracking the position of a moving object;

[0016]FIG. 2 is a schematic diagram showing two tone signals and theirrespective phases between a tag transmitter and a receiver of the systemshown in FIG. 1;

[0017]FIG. 3 is a block diagram showing the functional elements of thetag transmitter used in the first embodiment;

[0018]FIG. 4 is a timing diagram illustrating the way in which the tagshown in FIG. 3 outputs the two tone transmit signal;

[0019]FIG. 5 is a block diagram showing the functional elements of thereceiver used in the first embodiment;

[0020]FIG. 6 is a block diagram showing the functional elements of aposition processor used in the first embodiment to process the signalsreceived from all of the receivers to determine the current position ofthe moveable object;

[0021]FIG. 7 is a schematic diagram showing a tracking system of asecond embodiment for tracking the position of a moving object;

[0022]FIG. 8 is a graphical representation of the sampling process usedby the DSP of the receiver used in the second embodiment;

[0023]FIG. 9a is a block diagram showing the functional elements of adigital signal processor block which forms part of the receiver shown inFIG. 5;

[0024]FIG. 9b is a flow chart illustrating the main processing stepsperformed by the digital signal processor of the receiver shown in FIG.5;

[0025]FIG. 9c is a graphical representation of the different FFT resultsobtained for each tone of each chirp received by the receiver;

[0026]FIG. 10 is a block diagram showing the functional elements of theposition processor used in the second embodiment to determine thecurrent position of the moveable object;

[0027]FIG. 11 is a flow chart showing the main operational stepsperformed by the elements of the tracking system in a third embodiment;

[0028]FIG. 12 is a block diagram showing the functional elements of theposition processor of the third embodiment;

[0029]FIG. 13 shows a conceptual arrangement of a number of receiversaround a horse-racing track to receive locator chirps transmitted by themobile tags carried by each horse;

[0030]FIG. 14 is a block diagram showing the functional elements of atag transmitter used in a fourth embodiment;

[0031]FIGS. 15a and 15 b are time plots illustrating the form of signaltransmitted by the tag transmitter shown in FIG. 13;

[0032]FIGS. 16a, 16 b and 16 c are time plots illustrating the use ofthe relative wavelengths of the frequency differences to determine thecoarse, medium and fine position estimates;

[0033]FIG. 17 is a block diagram showing the functional elements of adigital signal processor which forms part of the receiver of the fourthembodiment;

[0034]FIG. 18 is a frequency plot illustrating two parts of the receivedsignal's spectrum that are processed by respective processing channelswhich form part of the digital signal processor shown in FIG. 17;

[0035]FIG. 19 is a block diagram showing the functional elements of theposition processor of the fourth embodiment;

[0036]FIG. 20 is a block diagram showing the functional elements of aphase difference tracking loop for tracking the difference in phase inthe position processor of the fourth embodiment;

[0037]FIG. 21 is a block diagram showing the functional elements of atag transmitter used in a sixth embodiment; and

[0038]FIG. 22 is a block diagram showing the functional elements of areceiver used in the sixth embodiment.

[0039] First Embodiment

[0040] Overview

[0041]FIG. 1 illustrates the tracking environment 1 in which thetracking system of the present embodiment operates. The tracking systemis used in this embodiment for tracking the position of a mobile tag 2which is attached to a jockey on a horse (not shown) which is to betracked. The mobile tag 2 carried by the jockey transmits a signal whichis received, in this embodiment, by three fixed receivers 3-1, 3-2 and3-3. The receivers 3 process the received signals and transmit theprocessed signals to a position processor 4 which then calculates theposition of the mobile tag 2 from the signals received from thereceivers 3. In this embodiment, the mobile tag 2 transmits two tones(tone A and tone B) of different frequency which enables the system tobe able to determine the absolute position of the mobile tag over arelatively large operating range whilst maintaining position sensingaccuracy. The reason for this will now be described with reference toFIG. 2.

[0042] When a single tone is transmitted between the tag 2 and areceiver 3, in order to be able to determine absolute position from ameasurement of the phase of the received signal, the distance betweenthe tag 2 and the receiver 3 must be less than the wavelength of thetransmitted tone. If this is not the case, then a phase ambiguityproblem arises. As discussed above in the introduction, for anapplication such as horse racing, this may require a transmissionfrequency in the kilohertz or megahertz part of the radio spectrum.However, when two tones of different frequencies are transmitted, it isthe difference between the frequencies which sets the maximum possibleunambiguous range of measurement. This is because, as illustrated inFIG. 2, the instantaneous phase relationship between the two tones (toneA and tone B) changes in each wavelength and repeats at a frequencygiven by the difference between the frequencies of the two tones.Therefore, with a two tone system, the maximum unambiguous range ofmeasurement is given by the following equation: $\begin{matrix}{{{MAX}\quad {RANGE}} = \frac{c}{f_{A} - f_{B}}} & (1)\end{matrix}$

[0043] where c is the speed of light, f_(A) is the frequency of tone Aand f_(B) is the frequency of tone B. In other words, the maximum rangeis not dependent on the actual frequency of the transmitted signals butonly on their difference in frequency. Therefore, frequencies from partsof the radio spectrum which are not regulated can be used. For example,two tones separated by 1 MHz could be transmitted within the 2.4 to2.485 GHz bandwidth which is allocated for use without a licence inaccordance with IEEE Standard 802.11. Such a system would be able toprovide absolute position measurement over a range of approximately 300metres (whereas a single tone at such a frequency would provide anunambiguous range of measurement of about 10 cm).

[0044] Mobile Tag

[0045] A description will now be given with reference to FIGS. 3 and 4of the functional elements of the mobile tag 2 used in the firstembodiment. As shown, the tag 2 has a Field Programable Gate Array(FPGA) 10 which receives a clock input from a crystal oscillator (CLK)11. The FPGA 10 outputs data identifying the frequency, the startingphase and the duration of a signal to be synthesised to a Direct DigitalSynthesizer (DDS) 12. In response, the DDS 12 generates the tone at thedesired frequency starting from the described start phase and for thedesired duration. In this embodiment, the FPGA 10 is programmed to causethe DDS 12 to generate a first frequency, followed by a secondfrequency, followed by a pause, followed once again by the firstfrequency then the second frequency and again a pause in a constantlyrepeating pattern. In this embodiment, the DDS 12 does not directlygenerate gigahertz signals. Instead, it generates intermediatefrequencies in the range of 70 MHz to enable the use of simplercomponents therein. The digital signal output by the DDS 12 is thenconverted into an analogue signal by the digital-to-analogue converter(DAC) 14. This signal is then up-converted to the appropriatetransmission frequency (2.410 GHz for tone A and 2.409 GHz for tone B inthis embodiment) by mixing it in a mixer 16 with an appropriate mixingsignal generated by the local oscillator 18. In this embodiment, thelocal oscillator 18 is programmable and generates a mixing frequency asdefined by a signal received from the FPGA 10. The mixed signal is thenfiltered by a filter 20 to remove unwanted components from the mixingoperation and then the filtered signal is amplified by a power amplifier22 and transmitted (broadcast) via the antenna 24.

[0046]FIG. 4 illustrates the form of the two tone signal transmitted bythe mobile tag 2 in this embodiment. As can be seen from FIG. 4, the twotones (tone A and tone B) have different frequencies (not shown toscale), with tone A being transmitted first then tone B followed by notone, followed by tone A again, tone B, no tone and so on. Each pulsesequence of tone A followed by tone B transmitted by the mobile tag 2will be referred to hereinafter as a chirp. In this embodiment, theduration of each tone pulse is approximately 300 μs giving a total chirpduration of approximately 600 μs and the chirp repetition interval isapproximately 100 ms.

[0047] The dashed lines shown in FIG. 4 illustrate that when a tonestarts to be transmitted after a pause, the phase of the tone at thattime is the same as it would have been had the tone been continuouslytransmitted since the last pulse. The start phase of each pulse isdetermined by the FPGA 10 from the start phase of the previous tone, thefrequency of the tone and the time elapsed since the beginning of thelast pulse. In this embodiment, the repetition rate is chosen to ensurethat each chirp starts from the zero phase point of a basic referencefrequency derived from the clock oscillator 11.

[0048] Since this embodiment determines the position of the mobile tag 2by considering the phase of a signal transmitted by the tag, it isimportant to consider the source of the signal generated by the tag, itsphase and any phase shifts added by the components in the tag. Thefundamental signal source in the tag is the crystal oscillator 11 usedto generate the system clock (CLK) which operates at some predeterminedfrequency (f_(clk)) and which has some initial phase (φ_(clk)(t)).Although not shown in FIG. 3, the DDS 12 generates its output using thisclock signal. It does this by, effectively, frequency multiplying theclock signal to generate the appropriate intermediate frequency signalswhich it outputs to the digital-to-analogue converter 14. Therefore, thephase of the signal output from the DDS 12 when tone A is transmittedcan be represented by: $\begin{matrix}{\varphi_{A}^{DDS} = {N_{A}{\varphi_{clk}(t)}}} & (2)\end{matrix}$

[0049] and the phase of the signal output by the DDS 12 when tone B istransmitted can be represented by: $\begin{matrix}{\varphi_{B}^{DDS} = {N_{B}{\varphi_{clk}(t)}}} & (3)\end{matrix}$

[0050] where N_(A) and N_(B) represent the effective multiple of theclock frequency for tone A and tone B respectively. The local oscillator18 operates in a similar manner so that the phase of the mixing signalcan be represented by:

φ_(LO)=Kφ_(clk)(t)  (4)

[0051] where K represents the effective multiple of the clock frequencyfor the mixing signal. The other components of the tag (i.e. the DAC 14,the mixer 16, the filter 20, the power amplifier 22 and the antenna 24)each introduce a phase delay. However, in this embodiment, it is assumedthat these phase delays are the same for each of the tones and thereforethe phase (φ_(A)) of the transmitted signal for tone A can berepresented by:

φ_(A) =N _(A)φ_(clk)(t)+Kφ _(clk)(t)+φ_(c)  (5)

[0052] and the phase (φ_(B)) of the transmitted signal for tone B can berepresented by:

φ_(B) =N _(B)φ_(clk)(t)+Kφ _(clk)(t)+φ_(c)  (6)

[0053] where φ_(c) is the constant phase delay added by the DAC 14, themixer 16, the filter 20, the power amplifier 22 and the antenna 24.

[0054] Receiver

[0055] The receivers 3-1, 3-2 and 3-3 used in this embodiment arefunctionally the same and a description of the functional elements ofone of the receivers 3 will now be given with reference to FIGS. 5, 6and 7.

[0056] As shown in FIG. 5, the signal is received by the receive antenna30 and is passed to a low noise amplifier 32 where the received signalis amplified. The amplified signal is then passed to a mixer 34 where itis mixed with a signal generated by local oscillator 36 to down-convertthe received signal from the transmitted gigahertz frequency to theintermediate frequency at approximately 70 MHz. As shown, the localoscillator 36 generates the mixing signal from a local clock signalwhich is generated from a crystal oscillator (CLK) 37 which is the sameas the oscillator 11 used in the mobile tag 2. The output from the mixer34 is then filtered by a bandpass filter 38 to remove unwanted frequencycomponents from the mixed signal and is then passed to ananalogue-to-digital converter 40 which converts the down-convertedsignals into digital signals. The digital samples output by the ADC 40are then input to a digital signal processor (DSP) 42 which processesthe samples to generate data that varies with the phase of the receivedsignal.

[0057] The signal received by the receiver 3 will correspond to thesignal transmitted by the mobile tag 2, however the passage of thesignal through the air introduces a further phase delay proportional tothe distance the signal has travelled. The received signal phase fortone A and tone B at receiver R can therefore be represented by:$\begin{matrix}{\varphi_{A}^{R} = {{N_{A}{\varphi_{clk}(t)}} + {K\quad {\varphi_{clk}(t)}} + \varphi_{c} + {\varphi_{dA}^{R}(t)}}} & (7)\end{matrix}$

[0058] and $\begin{matrix}{\varphi_{B}^{R} = {{N_{B}{\varphi_{clk}(t)}} + {K\quad {\varphi_{clk}(t)}} + \varphi_{c} + {\varphi_{dB}^{R}(t)}}} & (8)\end{matrix}$

[0059] In this embodiment, it is assumed that the crystal oscillators inthe receivers 3 and the mobile tag 2 are perfectly synchronised witheach other. Therefore, the terms of the received phase involvingφ_(clk)(t) can be ignored. Further, as with the similar components ofthe mobile tag 2, the receive antenna 30, the low noise amplifier 32,the mixer 34, the filter 38, the analogue-to-digital converter 40 andthe digital signal processor 42 will introduce a phase delay into thereceived phase. However, in this embodiment it is assumed that thesephase delays are constant for a given chirp and can be incorporatedwithin the expression for φ_(c).

[0060] The phase data generated by the DSP 42 is then passed, togetherwith a time stamp for the measurement and a receiver ID, to a datatransmitter 44 which, in this embodiment, packages the data using asuitable network protocol (such as TCP/IP) and transmits the data to theposition processor 4 over an appropriate data network. In the presentembodiment, the link between the receivers 3 and the position processor4 is made using a wireless network. That is a conventional computernetwork system implemented without wires but using radio transmitters.Examples of such a wireless network include AirPort™ and Wi-Fi™ systems.

[0061] Position Processor

[0062] Referring now to FIG. 6, the position processor 4 used in thisembodiment will now be described in detail. The data transmitted to theposition processor from all of the receivers 3 is received by datareceiver 70 which extracts the phase data from the network packaging andcontrol data that was added for transmission purposes. The extractedphase data is then passed to a measurement alignment unit 72 whichprocesses the received phase data to group the phase data for the samechirp from all of the receivers into a separate cluster. This isrequired since data transmitted over a TCP/IP network may not arrive atthe receiver in the order that it was transmitted. The measurementalignment unit 72 does this using the transmitted time stamp data and bywaiting until the data from all of the receivers for a given chirpshould have been received, allowing for the network latency.

[0063] The aligned measurements for a current chirp are then passed to aphase measurement determination unit 74 which performs a subtractionoperation to subtract the phase measurements associated with the tone Bsignal from the phase measurements associated with the tone A signal. Inparticular, the phase measurement determination unit 74 subtracts thephase measurement from receiver 1 for tone B from the phase measurementfrom receiver 1 for tone A, to generate a phase difference measurementfor receiver 1. The phase measurement determination unit 74 also doesthis for the phase measurements received from the other receivers. Inthis embodiment, there are three receivers 3-1, 3-2 and 3-3 whichreceive the tone A and tone B signals transmitted by the mobile tag 2.Therefore, the phase measurement determination unit 74 will generate thefollowing three phase difference signals, for each chirp transmittedfrom the tag, which are passed to the position determination unit 76.$\begin{matrix}\begin{matrix}{{{\Delta\varphi}^{1}(t)} = {{{\varphi_{dA}^{1}(t)} - {\varphi_{dB}^{1}(t)}} = {{{d_{1}(t)}\lbrack {f_{A} - f_{B}} \rbrack}/c}}} \\{{{\Delta\varphi}^{2}(t)} = {{{\varphi_{dA}^{2}(t)} - {\varphi_{dB}^{2}(t)}} = {{{d_{2}(t)}\lbrack {f_{A} - f_{B}} \rbrack}/c}}} \\{{{\Delta\varphi}^{3}(t)} = {{{\varphi_{dA}^{3}(t)} - {\varphi_{dB}^{3}(t)}} = {{{d_{3}(t)}\lbrack {f_{A} - f_{B}} \rbrack}/c}}}\end{matrix} & (9)\end{matrix}$

[0064] where d₁(t) is the distance between the mobile tag 2 and receiver3-1 at time t; d₂(t) is the distance between the mobile tag 2 and thereceiver 3-2 at time t; d₃(t) is the distance between the mobile tag 2and the receiver 3-3 at time t; f_(A) is the frequency of thetransmitted tone A; and f_(B) is the frequency of the transmitted toneB. As can be seen from equation (9), by taking the phase difference ofthe phase measurements from each receiver, the common phase delay(φ_(c)) introduced by the electronic components of the mobile tag 2 andthe receivers 3 has been removed from the calculation.

[0065] The position determination unit 76 uses the three phasedifference measurements obtained from the phase measurementdetermination unit 74 (together with the known transmission frequenciesof the mobile tag 2) to generate a value for the distance between themobile tag 2 and each of the receivers 3. From these distances, itdetermines the position of the mobile tag relative to the known positionof the receivers 3. This position measurement will be an absolutemeasurement, provided the mobile tag 2 is within one wavelength of thebeat frequency (f_(A)-f_(B)) of the transmitted tones. The way in whichthese calculations are done is well known to those skilled in the artand will not be described further here.

[0066] SECOND EMBODIMENT

[0067] In the first embodiment, it was assumed that the clocks in thetag and in the receivers 3 were synchronised to one another. Whilst thisis possible to achieve, it is impractical for most applications. Asecond embodiment will now be described in which the tag and thereceivers are not synchronised. In this embodiment, the mobile tag 2 hasthe same general architecture as the mobile tag 2 used in the firstembodiment. In this embodiment, a network calibration technique is usedto account for the lack of synchronisation between the receivers 3. Thiscalibration technique uses signals transmitted from a fixed tag 5 whoseposition is known and which is constructed and operates in the same wayas the mobile tag 2.

[0068] In this embodiment, the processing carried out by the digitalsignal processor 42 in each receiver 3 is different to the processingcarried out in the DSP 42 used in the first embodiment. A more detaileddescription will now be given of the operation of the ADC 40 and of theDSP 42 used in this embodiment with reference to FIGS. 8 and 9. In thisembodiment, the receivers 3 are arranged to digitise a frequency band of11 MHz which is centred around the 70 MHz intermediate frequency. Itdoes this using sub-sampling techniques by sampling the down-convertedsignal at 52 MHz. Sub-sampling this frequency band at this rate resultsin a digitised version of this 11 MHz band centred at 18 MHz. This isillustrated in FIG. 8 which shows the 11 MHz band 41 which is centred at70 MHz and the corresponding sub-sampled 11 MHz band 43 which is centredat 18 MHz. As shown in FIG. 8, this sub-sampled frequency band 43 liesentirely within the Nyquist band represented by the dashed box 45. Thetechniques of sub-sampling are well known and will not be describedfurther.

[0069] As shown in FIG. 9a, the samples generated by theanalogue-to-digital converter 40 are input to a digital mixing anddecimation unit 48 in the DSP 42, where the digitised frequency band 43is mixed to baseband to generate in phase (I) and quadrature phase (Q)samples which are then decimated by four (step S7-1 in FIG. 9b). Theresulting 13 mega I and Q samples per second are stored in a buffer 50.Blocks of these samples are then passed one block at a time to a FastFourier Transform (FFT) unit 52 which performs a complex FFT (step S7-3)using both the in phase (I) and quadrature phase (Q) signals in theblock. In this embodiment, the FFT takes a 256 point FFT on blocks of256 I and 256 Q samples. With the above sampling rate, this means thatthe FFT unit 52 produces an FFT output (which takes the form of an arrayof amplitude and phase values for a number of different frequencies foreach block of input samples) at a rate of one every 19.7 μs.

[0070] When the mobile tag 2 transmits a pulse either of tone A or toneB, the output from the FFT unit 52 should include an amplitude value anda phase value for that tone. Since the mobile tag 2 transmits pulses ofapproximately 300 μs of each tone, this means that there should be 15(300/19.7) consecutive FFT outputs having an amplitude and phase valuewhich corresponds to the transmitted tone. The FFT calculated for eachblock of samples is input to a signal comparison unit 54 whichdetermines whether or not the current FFT might form part of a chirp(step S7-5). It does this by comparing the amplitude values in thereceived FFT with an amplitude threshold stored in the store 56. Theresult of this comparison is passed to a control unit 58 which controlsthe position of a switch 60 so that if any of the amplitude values inthe current FFT are above the threshold, then those amplitude values andthe corresponding phase values are stored (step S7-7) together with anindication of the frequencies with which those amplitude values areassociated and with a time stamp identifying the current FFT. Theseamplitude and phase values will continue to be stored in the buffer 62until the signal comparison unit 54 and the control unit 58 identify(what they think is) the end of the chirp (step S7-9) by detecting whenthe amplitude values fall below the amplitude threshold 56.

[0071] As those skilled in the art will appreciate, whilst the use ofthe comparison unit and the amplitude threshold avoids the processing ofgeneral background noise, sometimes the background noise at particularfrequencies will be above the threshold and will cause the correspondingFFT values to be stored in the buffer 62. Therefore, in this embodiment,the data values stored in the buffer 62 are passed to a pattern matcher64 which looks for patterns in the data stored in the buffer 62 whichare characteristic of a chirp produced by the mobile tag 2. Inparticular, as mentioned above, the mobile tag 2 outputs a chirpcomprising approximately 300 μs of tone A followed by approximately 300μs of tone B. Therefore, the FFT data corresponding to a chirp shouldinclude an amplitude and phase value corresponding to tone A in fifteenconsecutive FFT outputs followed by an amplitude and phase valuecorresponding to tone B in fifteen consecutive FFT outputs. Thisexpected pattern is stored in the reference pattern store 66 and thepattern matching unit 64 compares the data stored in the buffer 62 withthis reference pattern in order to determine whether or not the dataactually corresponds to a chirp. By performing this pattern matchingoperation, the receiver reduces further the risk of outputting erroneousposition information.

[0072] When the pattern matching unit 64 identifies that the data storedin the buffer 62 corresponds to a chirp, it determines a time stamp forthe chirp from a receiver clock and determines the optimum timeslots forthe presence of each tone. In this embodiment, the receiver clock is asimple sample counter which is incremented by one for each block of 256samples received. This information is then passed to the control unit 58which then extracts phase information for both tone A and tone B fromthe identified values stored in the buffer 62 and outputs (step S7-15)this phase information from the DSP 42 to the data transmitter 44. Inthis embodiment, the control unit 58 outputs a single set of phasemeasurements for each of tone A and tone B for each chirp. However, asmentioned above, the buffer 62 will hold fifteen consecutive FFT outputshaving amplitude and phase values which correspond to each transmittedtone. If the clock frequencies of the tag 2 and the receiver 3 areperfectly synchronised and chosen so that each of the tones is centredwithin the corresponding FFT frequency bin, then the fifteen FFT phasevalues for each of the transmitted tones will remain constant. However,since the clock frequencies are not synchronised in this embodiment, thephase terms for these fifteen FFT outputs will be different.Fortunately, during a single chirp, it is unlikely that thesynchronisation between the mobile tag 2 and the receiver 3 will changeand therefore the change in the phase values between successive FFToutputs should be approximately the same. This is illustrated in FIG. 9cwhich shows the fifteen phase values obtained from fifteen consecutiveFFT outputs and the line 69 which best fits these points, the gradientof which depends upon the lack of synchronisation between thetransmitter and receiver clocks.

[0073] Consequently, in this embodiment, the control unit 58 determinesthe gradient of the best fit line 69 (using a least squares regressionalgorithm) and outputs this slope measurement (referred to hereinafteras the phase slope measurement φ_(s)) together with the phase valuemeasured from the best fit line 69 at a position corresponding to one ofthe fifteen FFT outputs (referred to hereinafter as the phase offsetmeasurement φ_(o)). It does not matter which one of the phase values isused as the phase offset measurement. However, in order to avoidpossible problems with phase offset measurements at the beginning andthe end of the pulse, in this embodiment, the control unit outputs thephase offset measurement (φ_(o)) of the best fit line 69 correspondingto the eighth FFT (i.e. the FFT obtained in the middle of the tonepulse). In this embodiment, the frequency of the two tones A and B havebeen chosen so that they will both appear at approximately the sameposition within the corresponding FFT frequency bin relative to thecentre of that bin. As a result, the phase slope measurement for tone Aand the phase slope measurement for tone B should be approximately thesame. However, in this embodiment, separate phase slope measurements(φ_(sA) and φ_(sB)) are taken and used to detect for corruption of thechirp data. These two phase measurements are then output to the datatransmitter 44 together with the time stamp for that chirp and thereceiver ID.

[0074] In addition to receiving the chirps from the mobile tag 2, thereceivers 3 also receive chirps from the fixed tag 5. The receiversprocess these chirps in the same way to generate corresponding phasemeasurements for the signals received from the fixed tag 5. As will bedescribed below, the phase measurements obtained from the fixed tag 5are used to correct for the lack of synchronisation of the receivers 3.

[0075] Referring now to FIG. 10, the position processor 4 of the secondembodiment will now be described in more detail. In the positionprocessor 4, the data receiver 70 and measurement alignment unit 72operate in the same way as described above with reference to FIG. 6 inthe first embodiment. The purpose of the phase measurement determinationunit 74 is to subtract the phase offset measurement for tone B of agiven chirp received at a given receiver from the phase offsetmeasurement for tone A for the same chirp received at the same receiver.However, as noted above, there is a constant drift in the measured phasecaused by the lack of synchronisation between the mobile tag clock andthe receiver clock (measured as the phase slope measurement φ_(s)) andas there are 15 FFT operations between the phase offset measurement(φ_(oA)) for tone A and the phase offset measurement (φ_(oB)) for toneB, the phase measurement determination unit 74 must add in a correctionbased on the phase slope measurements φ_(sA) and φ_(sB) in order toextrapolate these measurements to a common time. In this embodiment, thephase offset measurements are extrapolated to a point in time midwaybetween the times of the two tones being subtracted. To do this, thedetermination unit 74 multiplies the phase slope measurement for tone A(φ_(sA)) by 7.5 (since normalised units of time are used to determinethe phase slope measurement φ_(s) rather than seconds) and then addsthis to the phase offset measurement for tone A (φ_(oA)). Thedetermination unit 74 also multiplies the phase slope measurement fortone B (φ_(sB)) by 7.5 and then subtracts this from the phase offsetvalue measured for tone B (φ_(oB)). Thus the sum performed by the phasemeasurement determination unit 74 is as follows: $\begin{matrix}{{{\Delta\varphi}_{TR}( {t = C} )} = {{\varphi_{oA}^{TR}( {t = C} )} + {7.5{\varphi_{sA}^{TR}( {t = C} )}} - \lbrack {{\varphi_{oB}^{TR}( {t = C} )} - {7.5{\varphi_{sB}^{TR}( {t = C} )}}} \rbrack}} & (10)\end{matrix}$

[0076] which in this embodiment gives the phase difference measure fortag T from the signals received at receiver R at the time correspondingto the middle of the chirp (i.e. at t=C). As in the first embodiment,the phase difference calculated is equivalent to subtracting equation(8) from equation (7) but this time not ignoring the φ_(clk)(t) terms asfollows: $\begin{matrix}{{{\Delta\varphi}_{TR}(C)} = {{( {N_{A} - N_{B}} ){\varphi_{clk}^{TR}(C)}} + {\varphi_{dA}^{TR}(C)} - {\varphi_{dB}^{TR}(C)}}} & (11)\end{matrix}$

[0077] where φ_(clk) ^(TR)(c) is the difference between the clock phaseof the tag (T) and the clock phase of the receiver (R) at the timecorresponding to the middle of the chirp (ie φ_(clk) ^(tag) (C)−φ_(clk)^(R) (C)). As before, the constant phase lag φ_(c) has been cancelledtogether with the common term involving the up-converter multiple K.

[0078] In this embodiment, the phase difference measurements obtainedfrom chirps transmitted by the mobile tag 2 are output directly to theadder 80 and the phase difference measurements obtained from chirpstransmitted by the fixed tag 5 are output to a network calibration unit78 which calculates correction values to be added to the phasedifference measurements obtained from chirps transmitted by the mobiletag 2 in the adder 80. The phase difference measurements obtained forthe mobile tag 2 vary with the phase difference between the clockfrequency of the tag 2 and the clock frequency of the receiver fromwhich the measurement is derived. In this embodiment the calibrationunit 78 calculates correction values to be added to these phasedifference measurements in order to effectively reference themeasurements from all of the receivers 3 back to a single clock—that ofthe fixed tag 5, thereby removing their dependance on the differentphases of the receiver clocks. It does this by adding the followingcorrection value: $\begin{matrix}{{{Correction}\quad {{value}(R)}} = {{- ( {N_{A} - N_{B}} )}{\varphi_{clk}^{fxdR}( {t = C} )}}} & (12)\end{matrix}$

[0079] where φ_(clk) ^(fxdR)(C) represents the difference in the phaseof the fixed tag 5 relative to the phase of the receiver R at the timecorresponding to the middle of the chirp transmitted by the mobile tag 2(ie φ_(clk) ^(fxd)(t=C)−φ_(clk) ^(R) (t=C)). Since the position of thefixed tag is known, the value of φ_(clk) ^(fxdR) at the timecorresponding to when the fixed tag transmits its chirp can bedetermined. However, since there is likely to be a frequency offsetbetween the frequency of the clock in the fixed tag and the frequency ofthe clock in the receiver, this phase difference will have changed bythe time that the chirp from the mobile tag is received. Therefore, inthis embodiment, the network calibration unit 78 monitors the way inwhich φ_(clk) ^(fxdR) changes with time by monitoring how these valuechanges over a number of chirps transmitted by the fixed tag 5. It thenuses this history of information to determine what φ_(clk) ^(fxdR) willbe at the time of the chirp from the mobile tag. It then uses this valueto work out the appropriate correction value using equation (12) above.

[0080] Thus, when a phase difference value for a chirp transmitted bythe mobile tag 2 and received by receiver 3-1 is output by the phasemeasurement subtraction unit 74, calibration unit 78 outputs thespecific correction value for that chirp and for receiver 3-1, to theadder 80 where it is added to the phase difference measurement from thedetermination unit 74.

[0081] Adding the appropriate correction value to equation 11 gives thefollowing corrected phase difference measurement: $\begin{matrix}\begin{matrix}{{{\Delta\varphi}_{TR}^{corr}(C)} = {{\varphi_{oA}^{TR}(C)} - {\varphi_{oB}^{TR}(C)} + {15{\varphi_{s}^{TR}(C)}} - {( {N_{A} - N_{B}} ){\varphi_{clk}^{fxdR}(C)}}}} \\{= {{( {N_{A} - N_{B}} )\lbrack {{\varphi_{clk}^{mob}(C)} - {\varphi_{clk}^{fxd}(C)}} \rbrack} + {\varphi_{dA}^{TR}(C)} - {\varphi_{dB}^{TR}(C)}}}\end{matrix} & (13)\end{matrix}$

[0082] As can be seen from equation 13, the corrected phase differencevalues are no longer dependent on the phase of the receiver clocks.Instead they are all referenced back to the clock phase of the fixed tag(i.e. φ_(clk) ^(fxd) (C)). These corrected phase difference measurementsare then passed to the position determination unit 76 and used to solveequation 13 to find the position of the mobile tag 2 and to determinethe phase of the mobile tag's clock relative to that of the fixed tag 5(at the time of the current chirp being processed). In this embodiment,the position determination unit 76 uses an iterative numerical reductionmethod to solve for these unknowns from these corrected phase differencemeasurements. The way that it does this will now be described in moredetail. In order to illustrate the calculations that are performed bythe position determination unit 76, it is necessary to expand equation13 to introduce the distance between the mobile tag 2 and the respectivereceivers 3. The relationship between (φ_(dA) ^(TR)(C)−φ_(dB) ^(TR)(C))is given in equation 9 which can be expanded further in terms of theclock frequency of the tag 2 to give: $\begin{matrix}{{{\varphi_{dB}^{TR}(C)} - {\varphi_{dA}^{TR}(C)}} = {( {N_{A} - N_{B}} )f_{clk}{{d_{TR}(C)}/c}}} & (14)\end{matrix}$

[0083] Where f_(clk) is the frequency of the clock 11 of the mobile tag2. Substituting this into equation 13 gives: $\begin{matrix}{{\Delta \quad {\varphi_{TR}^{corr}(C)}} = {( {N_{A} - N_{B}} )\lbrack {{\varphi_{Tf}(C)} + {f_{clk}{{d_{TR}(C)}/c}}} \rbrack}} & (15)\end{matrix}$

[0084] where φ_(Tf)(C) is the phase of the mobile tag clock relative tothat of the fixed tag clock at the time of the current chirp (C). Theunknowns in this equation are φ_(Tf)(C) and d_(TR)(C). Since there arethree receivers, there will be three equations involving the fourunknowns φ_(Tf)(C), d_(T1)(C), d_(T2)(C) and d_(T3)(t). However, as thepositions of the receivers 3 are all known, the three distance measurescan be re-referenced relative to a common origin and written in terms ofa two dimensional position coordinate (d_(Tx)(t), d_(Ty)(t)) using thefollowing formula.

(d _(Tx)(t)−x_(R))²+(d _(Ty)(t)−y_(R))²=(d _(TR)(t))²  (16)

[0085] Where (x_(R),y_(R)) is the position of receiver R in terms ofthis coordinate system. Substituting the above into equation 15 gives:$\begin{matrix}{{\Delta \quad {\varphi_{TR}^{corr}(C)}} = {( {N_{A} - N_{B}} )\lbrack {{\varphi_{Tf}(C)} + {f_{clk}/{c\lbrack {( {{d_{Tx}(C)} - x_{R}} )^{2} + ( {{d_{Ty}(C)} - y_{R}} )^{2}} \rbrack}^{1/2}}} \rbrack}} & (17)\end{matrix}$

[0086] Therefore, there are now three unknowns (d_(Tx)(C), d_(Ty)(C),and φ_(Tf)(C)) and three measurements (Δφ_(TR) ^(corr)(C)), from whichthese unknowns can be calculated. As mentioned above, in thisembodiment, an iterative numerical reduction method is used to solve forthese unknowns. This is done by firstly defining, the function f_(i)(C)for each of the measurements (i) which equals the right hand side ofequation 17 minus the left hand side. This function f_(i)(C) should beequal to zero, however, due to approximations and other errors, it islikely that there will be a slight offset from zero. The positiondetermination unit 76 then finds the values of the unknowns whichminimise the sum of squares of these functions f_(i)(C), ie:$\begin{matrix}{{F( {d_{Tx},d_{Ty},\varphi_{Tf}} )} = {\sum\limits_{i = 1}^{3}\quad {f_{i}^{2}(C)}}} & (18)\end{matrix}$

[0087] As this is a continuous and differentiable function, a set ofpartial derivatives of F for d_(Tx)(C), d_(Ty)(C) and φ_(Tf)(C) arederived and the equation solved numerically. This is done using theBroyden-Fletcher-Goldfarb-Shanno method which is a variant of theDavidon-Fletcher-Powell algorithm. This is a standard minimisationalgorithm which finds the values of the unknown variables that minimiseF and therefore a further description of it shall be omitted. The readeris referred to the publication “Numerical recipes in C,” by Press,Teukolsky, Vettering and Flannery for further details of this algorithm.

[0088] Third Embodiment

[0089] In the first and second embodiments described above, the positionof a single mobile tag was determined and then tracked. A thirdembodiment will now be described in which there is more than one mobiletag 2 to be tracked. FIG. 11 is a schematic flow chart illustrating theoperation of this embodiment for tracking N tags simultaneously. At stepS11-1, tag 1 transmits a chirp. This chirp is received by receivers 1, 2and 3 at steps S11-3, S11-5 and S11-7 respectively. Each of thereceivers 1, 2 and 3 processes the chirp and transmits the phasemeasurement data to the position processor. In step S11-9, tag 2transmits a chirp. This chirp is received by receivers 1, 2 and 3 atsteps S11-11, S11-13 and S11-15 respectively. Again, the receiversprocess the received chirp and transmit the phase measurement data tothe position processor. This process continues until the last tag, tagN, transmits a chirp at step S11-17 which chirp is received by thereceivers 1, 2 and 3 at steps S11-19, S11-21 and S11-23 respectively.Thereafter tag 1 transmits another chirp followed by tag 2 etc. Asbefore, the receivers 1, 2 and 3 process each received chirp andtransmit the phase measurement data to the position processor 4. Whenthe position processor receives the phase measurements for a tag, itimmediately calculates the position and clock offset for that tag atstep S11-25.

[0090] As those skilled in the art will appreciate, provided that eachtag transmits on different frequencies, it is possible for all of thetags to transmit simultaneously. Alternatively, if frequencies are to beshared between the tags, then it is necessary for at least those tagssharing a frequency to transmit at different times. In this embodiment,however, each of the tags transmits on different frequencies so that thephase measurements received from the receivers can more easily beassociated with the tag that transmitted the chirp. In the alternativeembodiment where tags share frequencies, either the system must knowwhen each tag is transmitting, or it must be able to deduce this fromthe determined position and from the previous positions of the tags thatare sharing frequencies or some tag ID must be transmitted with thetones.

[0091] Referring now to FIG. 12, the functional elements of the positionprocessor 4 used in this embodiment will now be described in moredetail. The data receiver 70, the measurement alignment unit 72, thephase measurement subtraction unit 74, the network calibration unit 78,the adder 80 and the position determination unit 76 all operate in thesame way as the corresponding elements of the second embodimentdescribed above. However the output from the position determination unit76 is, in the present embodiment, output to a clock offset processingunit 82 and a path processing unit 84. The clock offset processing unit82 provides a feedback estimate of the phase of the mobile tag's clockrelative to that of the fixed tag (φ_(Tf)(C)) for each tag 2 to theposition determination unit 76, in order to speed up the minimisationalgorithm. In this embodiment, the clock offset processing unit 82calculates the feedback estimates by considering the history of therelative phase for a mobile tag and the fixed tag and extrapolating fromit to provide an estimated phase at the next chirp. This phase estimateis then used by the algorithms in the position determination unit 76 asa starting estimate for the relative phase (φ_(Tf)(C)) during theprocessing of the signals from the next chirp from that tag 2.

[0092] The path processing unit 84 applies certain physical rules to theposition data output by the position determination unit 76 to ensurethat the position solution does not alter in such a fashion that wouldimply a physically impossible movement of the tag 2. For example, if thetags are constrained to move over a predetermined course, then positionsoutside this course must be invalid and so those position solutions arenot allowed. The path processing unit 84 also uses time averaging todetermine velocity information for each tag 2 and thus the output fromthe path processing unit 84 is, in this embodiment, a position andvelocity for each mobile tag 2. As shown in FIG. 12, the output of thepath processing unit 84 is also fed back into the position determinationunit 76, also to provide starting estimates for the minimisationalgorithm for that tag at the next chirp. This estimate is determined,in this embodiment, using the determined velocity measurement and thetime between chirps from that tag.

[0093] Fourth Embodiment

[0094] Overview

[0095] A number of embodiments have been described above whichillustrate the way in which the present invention can be used todetermine the position of one or more moveable tags relative to a numberof receivers. A fourth embodiment will now be described with referenceto FIGS. 13 to 17 of a prototype system that has been built fordetermining and tracking the position of a number of horses around aracing track. FIG. 13 is a schematic diagram illustrating the racingtrack 199 and showing three horses 200-1, 200-2 and 200-3 withassociated riders 201-1, 201-2, 201-3 racing around the racing track199. Attached to each rider 201 is a tag 2 which is similar to themobile tag described in the above embodiments. In this embodiment, thereare four receivers 3-1, 3-2, 3-3 and 3-4 which receive the chirpstransmitted by the mobile tags 2. In this embodiment, there are also twofixed tags (not shown) which are the same as the fixed tags used in thesecond embodiment and used for the same purpose. FIG. 13 also shows achirp that is transmitted by tag 2-1. In this embodiment, the tags 2 arearranged to share transmission frequencies but the chirp repetition ratefor each tag is different in order to minimise collisions caused by twotags transmitting at the same frequency at the same time. In thisembodiment, each chirp also includes a tag ID frequency which is uniqueand used to ensure that the correct phase measurements are associatedwith the correct tags.

[0096] Tag

[0097]FIG. 14 is a schematic block diagram illustrating the mainfunctional components of the tags 2 carried by the riders 201. Asbefore, an FPGA 10 receives a clock input (which is in the presentembodiment is at 13 MHz) from the clock 11 and provides instructions toa DDS 12 to generate the required tone signals. As will be describedbelow with reference to FIG. 15a in this embodiment, each chirpcomprises a predetermined pattern of six different tones. The FPGA 10also receives data defining a tag ID frequency from the tag ID store 13.This tag ID data defines a unique ID frequency associated with theparticular tag 2. This tag ID data is also provided by the FPGA 10 tothe DDS 12 so that a tone with the frequency F_(ID) can be generated bythe DDS 12. The tones generated by the DDS 12 are generated at afrequency of approximately 70 MHz and require conversion into analoguesignals and mixing up to the transmission frequency at approximately2.45 GHz. In the present embodiment, this is achieved using the DAC 14and a two-stage mixing process using mixers 16 and 27. In thisembodiment, mixer 16 receives a mixing signal from a first localoscillator 18 whose frequency is also controlled by the FPGA 10. Themixer 16 up coverts the tones from the DDS 12 to an intermediatefrequency at approximately 450 MHz. The mixed signal is then filtered bythe bandpass filter 20 to remove unwanted frequency components of themixing operation and is then input to the second mixer 27. As shown, thesecond mixer 27 receives the mixing signal from a second localoscillator 26 whose frequency again is controlled by the FPGA 10. Thefrequency of the second mixing signal is such as to cause the tonesoutput from the DDS 12 to be mixed up to a frequency of approximately2.45 GHz. This signal is then filtered by the bandpass filter 28, againto remove unwanted frequency components from the mixing operation. Thefiltered signal is then amplified by the power amplifier 22 before beingtransmitted from the transmit antenna 24.

[0098] In the first embodiment described above, each chirp included twotones (tone A and tone B). The use of two tones in this way allowed thedetermination of phase difference measurements which increased the rangeover which an absolute position measurement could be obtained. As thoseskilled in the art will appreciate, it is possible to further improvethis system by introducing more tones into the chirp so that more tonedifferences can be calculated. The form of the chirps transmitted byeach of the tags 2 in this embodiment will now be described withreference to FIG. 15. In particular, FIG. 15a shows the tone pattern ofthe chirp, which is a sequence of seven tones. The chirp begins with atone at frequency f₀ which is transmitted for 1 ms. This initial part ofthe chirp is used a “warm-up” signal and is not used for positioncalculation. It is there to allow the components in the transmitter andthe receiver to warm-up in order to reduce signal degradation in thesubsequent tones. Following the transmission of the tone at frequencyf₀, four tones with frequencies f₁, f₂, f₃ and f₄ are transmitted insequence each for 0.3 ms, followed by another tone at frequency f₀ againfor 0.3 ms. In this embodiment, these four tones and the second burst ofthe f₀ tone are used for position calculations. Following the secondtone at frequency f₀, the ID tone (as up converted through the mixers)at a frequency of f_(ID) is transmitted. As mentioned above, the IDfrequency is unique for the respective tags 2 which allows the receivers(and/or the position processor) to identify the tag which transmittedthe current chirp phase measurements that are being processed.

[0099]FIG. 15b illustrates the spread of frequencies that aretransmitted over the tone. As shown, frequency f₁ is higher than f₀ andfrequencies f₂, f₃ and f₄ are lower than frequency f₀ by differingamounts. By considering the tone f₀ as a centre frequency around whichthe others are generated, the exact frequency differences between thesetones in this embodiment are: Tone Frequency relative to f₀ f₀ 0 MHzf₁ + 5.12890625 MHz f₂ − 0.1015625 MHz f₃ − 0.7109375 MHz f₄ −4.82421875 MHz f_(ID) unique for each tag

[0100] f_(ID) is generated in the present embodiment to be f₀ plus orminus 0 to 32 times 101.5625 kHz, yielding a maximum of 65 tags. Itshould be noted that all of the frequencies f₁ to f₄ and f_(ID) areinteger multiples of 50.78125 kHz which is used as a base frequency inthe tags 2 and the receivers 3. As mentioned above, the chirp repetitionintervals for each of the tags are different but are all approximately100 ms. The exact repetition rates are chosen to ensure that each chirpstarts from the zero phase point of the 50.78125 kHz basic referencefrequency discussed above. This basic reference frequency represents thegranularity of the frequency spacing for the tones within the chirp andis the basic “bin width” of the FFT used in the DSP 42 of the receiver 3for extracting the tone phases. The 50.78125 kHz base frequency isgenerated as {fraction (1/256)} of the 13 MHz clock oscillatorfrequency.

[0101] These frequency spacings allow the calculation of the followingfrequency differences between the tones: one difference of approximately0.1 MHz (f₀−f₂), two differences of approximately 0.7 MHz (f₂−f₃=0.6 MHzand f₀−f₃=0.7 MHz) and five differences of approximately 5 MHz(f₁−f₀=5.1 MHz, f₁−f₂=5.2 MHz, f₀−f₄=4.8 MHz, f₂−f₄=4.7 MHz andf₃−f₄=4.1 MHz). These phase differences allow a coarse positionmeasurement to be calculated using the 0.1 MHz phase differencemeasurements (which corresponds to a maximum unambiguous distance ofapproximately 3000 m), an intermediate position measurement to becalculated using the 0.7 MHz phase difference measurements (whichcorrespond to a maximum unambiguous distance of approximately 430 m) anda fine position measurement to be calculated using the 0.5 MHz phasedifference measurements (which corresponds to a maximum unambiguousdistance approximately 60 m). Referring to FIG. 16, in the presentembodiment, the position processor operates initially using only the 0.1MHz difference signal (illustrated in FIG. 16a) to obtain a coarseposition measurement. It then uses this coarse position measurement toidentify the correct phase cycle of the 0.7 MHz difference signal(illustrated in FIG. 16b) from which a medium accuracy measurement isdetermined. Finally, it uses this medium accuracy measurement toidentify the correct phase cycle of the 5 MHz difference signal(illustrated in FIG. 16c) from which a fine position measurement isdetermined.

[0102] Receiver

[0103] The receivers 3 used in this embodiment are substantially thesame as those used in the second embodiment described above. However,there are some differences in the structure of the analogue to digitalconverter and the digital signal processor that are used. Thesedifferences are mainly designed to ensure that the system can beoperated using a Pentium III PC compatible computer. FIG. 17 is aschematic block diagram illustrating the main components of the ADC 40and the DSP 42 used in this embodiment. As shown, the ADC 40 comprisestwo identical 12 bit ADCs 41 a and 41 b each of which receive the sameinput signal from the filter 38 (see FIG. 5). As in the secondembodiment, each of the ADCs 41 a and 41 b is configured to undersamplethe signal at 52 megasamples per second. This produces a signal imagecentred at 18 MHz. The output from the ADC 41 a is passed to DSP 42where it is fed to a first mixing and decimation unit 48 a and theoutput from ADC 41 b is passed to the DSP 42 where it is fed to a secondmixing and decimation unit 48 b. As shown in FIG. 17, the data streamfrom ADC 41 a is passed first into a complex digital local oscillator(DLO) 120 a which, in this embodiment, mixes the data stream with a15.4609375 MHz mixing signal. As it is a complex DLO, the output fromthe DLO 120 a comprises both in phase (I) and quadrature phase (Q)samples. Each of the (I) and (Q) sample streams are then low passfiltered by a respective low pass filter 122 a and 122 b which have a 1dB cut-off frequency of 5.2 MHz. The filtered I and Q data streams arethen decimated by eight down to a sample rate of 6.5 megasamples persecond by the respective decimator units 124 a and 124 b. The outputs ofthese decimators, which form the output from the mixing and decimationunit 48 a, are then passed into a respective buffer 50 a and 50 b.Blocks of both the in-phase and quadrature phase samples from thesebuffers are then input to an FFT unit 52 a which performs a complex FFTin the manner described above in the second embodiment. In thisembodiment, however, the FFT unit 52 a performs a 128 point complex FFTrather than a 256 point FFT.

[0104] The digital samples output from the ADC 41 b are passed to acomplex digital local oscillator 120 b which, in this embodiment, mixesthe data stream with a 20.5390625 MHz mixing signal. The output in-phaseand quadrature phase data streams are then low pass filtered by arespective low pass filter 122 c and 122 d, both of which have a 1 dBcut-off frequency of 5.2 MHz. The filtered I and Q data streams are thendecimated by eight down to a sample rate of 6.5 megasamples per secondby the decimator units 124 c and 124 d. The outputs from thesedecimators are then input to a respective buffer 50 c and 50 d. Again,blocks of 128 in-phase and quadrature phase samples from these buffersare then input to an FFT unit 52 b which performs a 128 point complexFFT on the samples in the block.

[0105] As those skilled in the art will appreciate, by mixing thesamples with different mixing frequencies by the DLOs 120 a and 120 b,different parts of the spectrum of the received signal are evaluated bythe two channels. With the sample rates used and the number of pointsconsidered in the FFT, this means that each frequency bin of the FFToutputs represents 50.78125 kHz of frequency spectrum, with the entireFFT output from the FFT unit 52 a representing the lower 6.5 MHz of thereceived signal spectrum and the output of the FFT unit 52 brepresenting the upper 6.5 MHz of the received signal spectrum. Theparts of the spectrum that are processed by the two channels areillustrated in FIG. 18. The dashed plot 121 illustrates the part of thesignal spectrum that is analysed by the FFT unit 52 a and the plot 123illustrates the part of the signal spectrum that is analysed by the FFTunit 52 b. The sloping off of the ends of these plots illustrate theeffects of the cut-off rate of the low-pass filters 122 used in therespective channels. As illustrated by the hatched area 125, there is anoverlap region centred at 18 MHz (which corresponds to the f₀ frequencytone). The location of the other tone signals within the chirp are alsoshown in FIG. 18 for information. In this embodiment, the mixingfrequencies have been chosen so that the frequency bins match in theoverlap region 125 so that they can be merged together into a single FFTarray spanning the desired range of frequencies for the tag chirp. Theresult is similar to what would have been achieved using a singleprocessing channel operating at 13 megasamples per second and using anFFT unit that carries out a 256 point FFT.

[0106] As shown in FIG. 17, the output from the FFT units 52 a and 52 bare input to the signal comparison unit 54 where the amplitude values ofthe FFTs are compared with the amplitude threshold 56 in order to detectthe beginning of a chirp. In this embodiment, this is done by detectingthe presence of a signal in the FFT output which corresponds to the f₀frequency tone which is transmitted at the beginning of each chirp. Whenthe beginning of a chirp is detected in this way, the amplitude signalsin each FFT frequency bin corresponding with known tone frequencies areused to construct a matrix having 5 rows (one for each tone frequency)and 180 columns (for 180 consecutive FFT outputs, which corresponds toapproximately 3.5 ms of received signal) which is sufficient to span anentire chirp. The pattern matching unit 64 then compares this pattern ofFFT values stored in the buffer 62 with the reference pattern 66 whichrepresents an ideal chirp response. This ideal chirp response is similarto the tone pattern shown in FIG. 15b. However it is not exactly thesame since, in this embodiment, the frequency of tone f₀ lies within theoverlap region 125 of the two FFTs. Therefore, when tone f₀ is beingtransmitted, the output from both of the FFT units 52 a and 52 b willinclude amplitude and phase values corresponding to that tone. Further,as shown in FIG. 18, tone f₂ lies just outside the region 125 and willnot be significantly attenuated by the low pass filters 122. Therefore,tone f₂ will also be represented in the output from both FFT units 52 aand 52 b. However, this is easily represented within the referencepattern and does not pose a problem to the pattern matching unit 64.

[0107] In this embodiment, the pattern matching unit 64 compares thepattern of FFT values stored in the buffer 62 by cross-correlating thereference pattern with the data in the buffer 62. This identifies thetime offset of the chirp within the sample set, and this time offset isused to determine the time base for the chirp in terms of the receiver'sclock. This time offset is also used to determine the optimum time slotsfor the presence of each tone within the data in the buffer 62. Once achirp has been identified within the data stored in the buffer 62, thecontrol unit 58 determines the tag ID from the received f_(ID) frequencyand extracts an amplitude measurement, a phase offset measurement and aphase slope measurement for the other tones in the chirp. Further, inthis embodiment, the control unit 58 determines two sets of amplitude,phase offset and phase slope measurements for the f₀ tone, one from thedata received from each of the two FFT units 52 a and 52 b. This ispossible, since the f₀ frequency appears in the spectrum of the receivedsignal which corresponds to the usable overlap region 125 from theoutputs of the FFT units 52. Similarly, two sets of measurements couldhave been obtained for the f₂ tone. However, this was not done in thisembodiment.

[0108] These amplitude, phase offset and phase slope measurements arethen transmitted from the receiver to the position processor togetherwith data identifying the receiver ID, the receiver time for the chirpand the tag ID. As in the embodiments described above, this message istransmitted via a wireless network to the position processor 4 as soonas it has been calculated.

[0109] In this embodiment, each receiver 3 is arranged to operate inthree different modes, with the mode being selected by the receiveraccording to the circumstances at that time. The three modes are a scanmode, a collect mode, and a refresh mode.

[0110] In the scan mode, the output from one of the FFT units 52 isprocessed by the signal comparison unit 54. In this embodiment, thisprocessing involves checking the frequency bin of the FFT outputcorresponding to the f₀ frequency for the presence of a signal. This isdetermined by comparing the amplitude value for the corresponding FFTbin against the fixed threshold which needs to be exceeded for apredetermined number (in this embodiment 5) of consecutive FFT outputs.When this occurs, the receiver is switched to the collect mode.

[0111] In the collect mode, the second processing channel is activatedso that both channels are working in parallel to process the receiveddata as described above with reference to FIGS. 17 and 18. During thecollect mode, the frequency bins for the relevant tones are stacked intothe buffer 62. As discussed above, this continues for 180 FFT outputs(corresponding to approximately 3.5 milliseconds of transmitted signal)which is enough to capture all of the transmitted chirp. This data isthen processed to extract the amplitude, phase offset and phase slopevalues as discussed above and then the operating mode of the receiver isswitched to the refresh mode.

[0112] In the refresh mode, the receiver operates in exactly the sameway as in the scan mode except that it is waiting for the absence of thesignal at the f₀ frequency, at which point it returns to the scan modediscussed above.

[0113] Position Processor

[0114] The operation of the position processor 4 used in this embodimentwill now be described with reference to FIGS. 19 and 20. The datareceived from each receiver 3 is received by the data receiver 70 andpassed to the measurement alignment unit 72 as before. The received datais also stored in a data store 71 for subsequent retrieval andprocessing. Storing the data in this way allows the system to reprocessthe data off-line which can be used to debug the system and forcomparative testing for algorithm development. In the measurementalignment unit 72, the incoming data packets, each containing the dataof a single tag chirp from one receiver, are queued in a first in, timesequenced out queue. The time sequencing is based on the receiver timetags appended to the chirp data. Since each receiver has its ownasynchronous clock, these time tags are referenced to the positionprocessor's clock using a clock difference derived statistically from alarge number of received packets. This statistically derived clockoffset is not used in the position processing algorithms but it isneeded to determine the association between chirps received at thedifferent receivers. It only needs to have an error smaller than halfthe minimum chirp interval which in this embodiment is approximately 46ms. The chirps are then drawn out from this queue in time sequence andpassed to a quality assessment (QA) and collision detection unit 73 viaa set of chirp smoothing filters (not shown).

[0115] The chirp smoothing filters are used to smooth out variations inthe determined phase slope measurements for each of the tones. Arespective smoothing filter is provided to smooth the chirp data fromeach receiver for each tone from each tag. Therefore, in thisembodiment, there are a hundred (5 tones×4 receivers×5 tags) chirpsmoothing filters. Smoothing is done since the phase slope measurementsfor a tone should not change significantly from one chirp to the next.Therefore, in this embodiment, each chirp smoothing filter performs arunning average calculation over a predetermined length of time on thecorresponding phase slope measurements. In this embodiment, the chirpsmoothing filters associated with the fixed tags 5 carry out a runningaverage over approximately one hundred seconds worth of chirps and thechirp smoothing filters associated with the mobile tags carry out arunning average over approximately ten seconds worth of chirps. Thesmoothed phase slope measurements output from these chirp smoothingfilters are then used in the subsequent analysis.

[0116] The QA and collision detection unit 73 operates to identifycollisions (ie when two tags are transmitting at the same time) and todiscard the chirp data when this occurs. In this embodiment, this isdone using knowledge about the chirp repetition rates of each tag. Inparticular, the QA and collision detection unit 73 monitors the chirprepetition rates of each tag and each time a reported chirp is received,the QA and collision detection unit 73 checks whether any two tags werescheduled to transmit at that time. If they are then the data for thatchirp is automatically discarded. The chirp data is also subjected to aset of consistency checks that test the amplitude and phase slopemeasurements for variation from one chirp to the next. In particular, ifthese values change significantly from one chirp to the next or if thephase slope measurements for a single chirp differ substantially, thenagain the data for that chirp is discarded. In this embodiment, the QAand collision detection unit 73 also compares the received tag IDsagainst a list of allowed tags and the received data for the chirp isdiscarded if the tag ID is not on this list.

[0117] The chirp data that is not discarded by the QA and collisiondetection rate 73 is then passed to the phase measurement subtractionunit 74 where the following phase subtraction measurements arecalculated: Phase difference measurements Beat frequency Δφ₀ f₀ − f₂ =0.1 Mhz (from channel 1 of the ADC) Δφ₁ f₀ − f₂ = 0.1 Mhz (from channel2 of the ADC) Δφ₂ f₂ − f₃ = 0.6 Mhz Δφ₃ f₀ − f₃ = 0.7 Mhz Δφ₄ f₁ − f₀ =5.1 Mhz Δφ₅ f₁ − f₂ = 5.2 Mhz Δφ₆ f₀ − f₄ = 4.8 Mhz Δφ₇ f₂ − f₄ = 4.7Mhz Δφ₈ f₃ − f₄ = 4.1 Mhz

[0118] Each phase difference measurement is calculated by referring thetwo tone phase offsets concerned to the time point between the two tonesusing the phase slope measurements for the chirp to extrapolate to thecommon time, and then subtracting them. As in the second embodiment,the-phase offset measurement for each tone is taken at a timecorresponding to the middle of the tone and this value is extrapolatedusing the associated phase slope measurement to the point in time midwaybetween the two tones being subtracted. For example, referring to FIG.15b, in the case of the subtraction the phase measurements for f₀ and f₃this time point lies somewhere in the middle of the tone at frequencyf₄. As mentioned above it is these extrapolated values (which representwhat the expected tone's phases would be at the same point in time) thatare subtracted. These phase differences are represented as an absolutephase value at the measurement time and a phase slope. This phase slopeis initialised by subtracting the two phase slope measurements for thetwo tones being subtracted and is thereafter maintained by a phaselocked loop which tracks the phase difference between chirps. Further,since the difference frequencies may undergo several cycles of phaserotation between chirps (depending on the relative clock frequencyoffsets between the tag and the receiver), the phase differencemeasurement is tracked between chirps.

[0119]FIG. 20 is a schematic block diagram illustrating the form of thephase difference tracking loop used in this embodiment. The loop isessentially a proportional and integral tracking control loop. The loopmaintains estimators of the phase difference offset value (Δφ_(o)^(A-B)) output from block 205 and of the phase difference slope value(Δφ_(s) ^(A-B)) output from the block 203. The estimators operate eachtime data for the corresponding chirp is received and at that time, theestimator values are updated. As shown, upon receipt of new phase offsetmeasurements for the two tones (labelled A and B), these are differencedin the adder 205. The current phase difference offset value from theestimator block 201 is then subtracted from this value in the adder 207to provide an error value (ξ). This error value then passes through theloop gain 209 and the low pass filter 211. The filtered error signal isthen used to update the phase difference slope value stored in the block203. As shown in FIG. 20, it does this by passing the error signalthrough a second amplifier block 213 and then subtracting from thisvalue, in the adder 215, the value of the previous phase differenceslope value provided by the delay unit 217.

[0120] This new phase difference slope value is then used to update thephase difference offset value stored in the block 201. It does thisfirstly by multiplying the new phase difference slope value in themultiplier 219 with the time between the last chirp and the currentchirp, which is provided by the chirp interval unit 221. This value isthen added together with a further amplified version of the error signaloutput from the amplifier 223 and the previous value of the phasedifference offset value provided via the delay unit 225. AS shown, thesevalues are added in the adder 227. This new value of the phasedifference offset value is then stored in the block 201 for use at thenext chirp time.

[0121] As shown in FIG. 20, this new phase offset value is also outputon the line 231 for use in the position calculation algorithms discussedin more detail below. Once this loop has locked onto the signals, it canalso be used to provide an estimate of the phase difference offset at anarbitrary time (τ) and not Just at the chirp times. As shown, this isachieved by multiplying the current estimate of the phase differenceslope value obtained from block 203 with the time (τ) in the multiplier235 and then by adding this to the current estimate of the phasedifference offset value output from the block 201 in the adder 237.

[0122] As those skilled in the art will appreciate, a separate phaselocked loop (PLL) is provided for each phase difference measurement thatis calculated, for each tag and for each receiver. Therefore, in thisembodiment, with nine phase differences, three mobile tags, two fixedtags and four receivers, this means there are 180 phase locked loopslike the one shown in FIG. 20.

[0123] System Calibration

[0124] As mentioned above, the receivers 3 operate independently of eachother and they each have their own unsynchronised clock. As in thesecond and third embodiments described above, the position processor 4uses the phase difference measurements obtained from the fixed receiversto reference the phase measurements from the mobile tags 2 back to asingle reference clock. In this embodiment, each fixed tag and eachmeasured phase difference for a mobile tag is treated independently sothat, in this embodiment, there are two independent reference clocks anddifferent phase measurements associated with each.

[0125] For each fixed tag (M) and each phase difference (P) a set ofφ^(MRP) (t) values is obtained from the corresponding phase differencetracking loops. Since the positions of the fixed tags and the receivers3 are known the phase rotation caused by the signal propagation pathsbetween the fixed tags and the receivers can be subtracted from thesephase difference measurements. This results in a set of modifiedφ′^(MRP) values for the fixed tags M, receivers R and phase differencesP, as though the fixed tags were located at each receiver. Bysubtracting these phase values from the corresponding phase differencesmeasured from a mobile tag, a phase measurement relative to the fixedtag is derived thereby eliminating the clock effects of the receivers.Again, the phase of each of these modified φ′^(MRP) values is trackedusing a separate phase lock loop (not shown) in order to estimate theirmost likely values at the time of the current position computation for amobile tag.

[0126] Position Calculation

[0127] In this embodiment, the position calculation is performed in asimilar manner to the way in which it was performed in the secondembodiment described above except using the phase difference valuesoutput from the phase difference tracking loops (one of which is shownin FIG. 19). Equation 18 given above for F is for a single fixed tag,one phase difference measurement and one mobile tag. Extending it to Mfixed tags and P phase difference measurements results in the followingfunction: $\begin{matrix}{{F( {d_{x},d_{y},\varphi_{{TF}_{pm}},t} )} = {\sum\limits_{m = 1}^{M}\quad {\sum\limits_{p = 1}^{P}\quad {\sum\limits_{i = 1}^{R}\quad {k_{pm}{f_{ipm}^{2}(t)}}}}}} & (19)\end{matrix}$

[0128] The value k_(pm) is a weighting factor that allows the differentpartial sums for different phase measurements and/or fixed tags to beweighted. For example, phase differences corresponding to longerwavelengths may be weighted lower than those associated with the shorterwavelengths, in order to balance the error each contributes. Again, thisfunction can be solved numerically to find best estimates of the valuesthat minimise F given the received measurements.

[0129] In this embodiment, there are two fixed tags, three mobile tags,four receivers and nine phase difference measurements being measured.Therefore, this results in 180 ((2+3)×9×4) individual phase differencemeasurements. For each mobile tag a set of 72 (2×4×9) phase differencemeasurements are obtained and there are 18 unknowns−16 unknown clockoffsets (φ_(TFpm)) and a two dimensional position. This set of equationstherefore contains significant redundancy (more equations thanunknowns). However, using additional fixed tags and phase differencemeasurements has been shown to yield significantly improved robustnessand accuracy through spatial diversity, frequency and time diversity andstatistical averaging of measurement noise.

[0130] Resolving Cyclic Ambiguity

[0131] In any phase measuring system there is a cyclic ambiguity thatcan result in a displacement error of integer multiples of wavelengths.In this embodiment, the short wavelength difference signals are around 5MHz which corresponds to a wavelength of approximately 60 metres, whichmeans that there is scope of many cycles of ambiguity in themeasurement. For this reason, the longer wavelengths are used to resolvethe cycle ambiguities. In principle, the long wavelength is used toproduce an unambiguous position within the area of coverage and havingan error small enough to initialise the medium wavelengths. Theseproduce a more accurate position in the region of the long wavelengthestimate and accurate enough to initialise the short wavelengths. Thealgorithm is then run using the short wavelengths to determine a highlyaccurate position fix.

[0132] Once a position fix has been obtained using the shortwavelengths, this position is used to determine an estimate for theposition calculation at the next chirp measurement, without having torestart the sequence through the long and medium wavelength steps.Referring to FIG. 18, the clock offset processing unit 82 and the pathprocessing unit 84 are used to provide these estimates for the positioncalculation for the next chirp. These operate in the same way as thecorresponding components in the third embodiment described above.

[0133] Once the position processor 4 is in the tracking mode, it stillcontinuously calculates the positions using the long and mediumwavelength measurements as well. The output from the position processor4 is taken from the short wavelength measurements, unless it isindicated as being invalid. In particular, the position measurementsoutput for the different wavelength measurements are continuallycompared in order to sense gross errors. If an error occurs, then theposition determination unit will detect this and correct for it bydiscarding the position from the shorter wavelength measurements.

[0134] Even when operating in the tracking mode it is still possible,for example because of fast motion of the tags, for there to be an errorin the cycle count for one or more of the phase difference measurements.Therefore, in this embodiment, the position processor 4 performs aseries of tests and iterations before arriving at the “best” positionsolution. In particular, the position processor 4 performs the followingprocessing steps for each positioning update using the short wavelengthmeasurements:

[0135] (i) Construct a matrix of R×M×P measurements from the measuredphase differences plus the calibration phase differences (where R is thenumber of receivers, M is the number of fixed tags and P is the numberof phase difference measurements corresponding to the short wavelengths(which in this embodiment is 5)).

[0136] (ii) Feed each of the R×M×P measurements through a respectivephase locked loop (similar to the PLL shown in FIG. 20), the output ofwhich is a smoothed “phase estimator” which is used in the positioncalculation.

[0137] (iii) Feed the set of M×P network phases (φ_(TFpn)) through arespective phase locked loop (again similar to the PLL shown in FIG.20), the outputs of which are used to estimate their most likely valuesat the time of the current position computation.

[0138] (iv) Based on the last known position for the tag, and hence thepath distances in wavelengths between the tag and each receiver, theestimated network phases (φ_(TFpn)) and the measured phases, determine amatrix of R×M×P cycle counts.

[0139] (v) Run the minimisation algorithm to derive the best fitposition (d_(x), d_(y)) and network phases (φ_(TFpn)) as well as theoverall function value of function F; and determine a matrix of errorresiduals representing the error contribution of all of the individualf_(ipm) equations.

[0140] (vi) Using the values of d_(x), d_(y) and φ_(TFpm) obtained fromstep (v), calculate new range phases and a new matrix of cycle counts.(The range phase is the phase value that is measured corresponding tothe distance between the receiver and the tag ignoring the cycle count.In particular, given a signal of wavelength λ and distance between thereceiver and the tag of d, the phase comprises the cycle count which isthe integer part of d/λ and the range phase which is the fractional partof d/λ).

[0141] (vii) Return to step (v) until the obtained function value forfunction F is equal to or greater than the previous value at which pointfurther minimisation is not being achieved.

[0142] (viii) Examine the individual error residuals matrix (i.e. theindividual values of f_(ipm)) to find the largest residual error andincrement or decrement the cycle count corresponding to that measurementset depending on the sign of the error.

[0143] (ix) Return to step (v) until the obtained function value forfunction F is equal to or greater than the previous value at which pointno further minimisation is being achieved.

[0144] (x) Disable the measurement set corresponding with the worstremaining residual error and then return to step (v) a small(configurable) number of times, to eliminate the worst few phase pathsfrom the position calculation.

[0145] The output of this process is then used in this embodiment as the“best” estimate of the mobile tag position and the network phase valuesfor the set of measurements.

[0146] Initialisation

[0147] The position processor 4 goes through a series of stages from theinitial start up to full tracking mode. In this embodiment, thesevarious stages are controlled by interlocking state machines running foreach tag and for the system state as whole.

[0148] The first state machine is for system start up and calibration.It runs independent processing calculations for each fixed tag andreaches the final system calibrated stage only when the required numberof fixed tags have reached this state. The processing for each fixed tagis as follows:

[0149] (i) Determine the initial chirp phase offset and slope values bydirect measurement.

[0150] (ii) Allow the chirp smoothing filter (used to smooth the phaseslope measurements) to settle.

[0151] (iii) Initialise the chirp phase difference tracking PLLs (shownin FIG. 20).

[0152] (iv) Allow the chirp phase difference tracking PLLs to settle.

[0153] (v) Wait for a required number of receivers to acquire lock thenset the tag synchronisation flag.

[0154] (vi) Wait for the required number of fixed tags to signalsynchronisation before moving to step (vii).

[0155] (vii) Set the system calibrated status once time synchronisationhas been achieved.

[0156] The system start up state machine takes several minutes toinitialise. This allows time for the receivers to stabilise and forstatistical time synchronisation of the receivers to be achieved. Alsothe filter and phase locked loop time constants for the fixed tags arenormally quite long compared to those for the mobile tags.

[0157] The second state machine is for mobile tag initialisation andposition processing. This operates as follows:

[0158] (i) Initialise the phase offset and slope values by directmeasurement.

[0159] (ii) Allow the chirp smoothing filter to settle.

[0160] (iii) Initialise the chirp phase difference tracking PLLs.

[0161] (iv) Allow the chirp phase difference tracking PLLs to settle.

[0162] (v) Wait for the required number of receivers to acquire lock.

[0163] (vi) Wait for system calibrated status.

[0164] (vii) Compute the initial tag position using the longestwavelength inputting the determined position into a position smoothingfilter (not shown) associated with the long wavelength and wait for thisfilter to settle.

[0165] (viii) Compute the tag position using the medium wavelengthinitialised by the position from the long wavelength and feed theresults into a medium wavelength position smoothing filter (not shown)and wait for this filter to settle.

[0166] (ix) Compute the tag position using the short wavelengthsinitialised by the position from the medium wavelength. Search thecyclic ambiguities for the best solution. If the function error residualtest shows an acceptable solution move to tracking mode.

[0167] (x) Track the position using the short wavelengths. Run themedium and long wavelengths in parallel to test for cycle jump errorconditions.

[0168] The filter and tracking time constants for the mobile tags arequite short and thus it is possible for this state machine to advanceall the way to full tracking mode in as little as 10 seconds in thisembodiment.

[0169] Although the above processes are-described as being “one-way”,they can be restarted under a number of error conditions. For example,if a chirp from one tag is not received for a predetermined length oftime (such as 10 seconds), its state machine can be reset appropriately.Similarly, if the short wavelength is found to be tracking with a cycleoffset, the position can be re-initialised starting with the medium orlonger wavelengths depending on the severity of the error.

[0170] Returning to FIG. 18, the final position determinationsdetermined by the path processing unit 84 are output to a motionfitting/time alignment unit 150. This unit allows tag positions for eachof the mobile tags 2 to be calculated for any arbitrary time, ratherthan the specific time at which the tag transmitted its chirp. It isnecessary to time align the position data especially since a gallopinghorse can cover approximately 1.7 metres between chirps. The smoothingand motion algorithms used by the unit 150 apply a least squaresstraight line fitting algorithm to the determined x and y positions overthe past few seconds worth of data. Time aligned sets of position datafor all of the mobile tags 2 are then extracted on a predetermined timebase, using the straight line fit parameters. In this embodiment, thisposition data is then transmitted over the Internet 152 to a remote racesimulation unit 154. The data is also stored in a data store 153 so thatit can be used subsequently for simulation purposes. In this embodiment,the remote race simulation unit 154 uses a graphical visualisation toolor a 3D game rendering tool which can generate an appropriate simulationof the race from the received position data.

[0171] Fifth Embodiment

[0172] In the fourth embodiment, all of the tags transmitted on the samefrequency but at different times and with the chirp repetition rate ofthe tags being different in order to minimise the times at which twotags will transmit at one time. However, as those skilled in the artwill appreciate, even with this approach, there is a limit to the numberof tags that can be operated within a given bandwidth. An embodimentwill now be described in which the centre frequency of the transmittedtones (ie the frequency value of tone f₀) is varied from one chirp tothe next in a predetermined manner. This also allows more tags to beoperated within a given bandwidth being processed. In particular, if thecentre frequency f₀ is frequency hopped in a pseudo random fashion fromone chirp to the next, then the probability of two tags transmitting atthe same frequency at the same time is very small and therefore a largernumber of tags can be tracked.

[0173] In such an embodiment, both the mobile tags and the fixed tagswould be arranged so that the FPGA 10 is programmed with a knowntransmit scheme which defines the centre frequency for each chirptransmitted from the tag. Thus the exact values of the frequencies f₀ tof₄ and f_(ID) will change for each chirp. However, the relationshipbetween each of the tones f₀ to f₄ and f_(ID) will remain fixed. In thereceivers, each receiver will know which frequencies are capable ofserving as the centre frequency f₀ according to the predeterminedtransmits schemes that are being used. Therefore, in the scan mode ofoperation, the receivers will scan all of the possible f₀ frequenciessimultaneously. The collect mode and the refresh mode for each receiverwill then work in the same way as in the fourth embodiment describedabove. In such an embodiment, the receiver would also transmit data tothe position processor 4 which identifies the centre frequency f₀ of thereceived chirp.

[0174] Upon receipt of the data from the receivers at the positionprocessor 4, the QA and collision detection unit 73 can monitor forcollision detections using the known transmit schemes for each of thetags. In particular, by comparing the transmit scheme for each tagagainst the transmits schemes of the other tags, based on the recentlyreceived chirps for each tag, the QA and collision detection unit 73 canlook ahead and predict when collisions can be expected. When the datafor these chirps are received they can then be discarded.

[0175] The use of such frequency hopping schemes in the tags alsoreduces the system's susceptibility to narrow band interference. Inparticular, in the fourth embodiment described above, if there is asource of interference over the transmission frequencies being used,then all of the chirp data is likely to be corrupted by noise. However,by frequency hopping the system's susceptibility to interference isreduced.

[0176] Sixth Embodiment

[0177] As a further alternative to frequency hopping, each of the tagsmay be arranged to transmit a spread spectrum signal rather than simpletones. A sixth embodiment will now be described which uses tags whichtransmits spread spectrum signals.

[0178]FIG. 21 is a functional block diagram illustrating the maincomponents of the tag 2 used in this embodiment. As shown, the tagincludes a signal generator 90 which receives a clock input from acrystal oscillator (not shown). In response to the clock input, thesignal generator 90 generates two tones corresponding to tones A and Bof the first three embodiments. The signal generator 90 also generates acontrol signal which it outputs to a pseudo-random noise (PN) codegenerator 92. The PN code generator 92 generates a pseudo-noise codewhich it outputs to a mixer 91 where the code is mixed with the tones Aand B to form the spread spectrum signal. The output from the mixer 91is then passed to a bandpass filter 94 and onto a power amplifier 96before being passed to a transmitter antenna 98 for transmission fromthe tag 2. In this embodiment, the frequency of the two tones A and Boutput from the signal generator 90 are sufficiently high to allow fordirect transmission. However, in an alternative embodiment, the outputfrom the mixer may be up converted to the appropriate transmissionfrequency.

[0179]FIG. 22 is a schematic block diagram illustrating the maincomponents of a receiver 3 used in such an embodiment. As shown, thesignals transmitted from the tag 2 are received by the receiver antenna100. The received signals are then amplified by a low noise amplifier102 and then down converted to an appropriate intermediate frequency inthe mixer 104. As shown, the mixer 104 receives the mixing signal from alocal oscillator 106 which receives a clock input from a crystaloscillator (not shown). The down converted signal output from the mixer104 is then passed through a bandpass filter 108 and then into across-correlator 110 where the received signal is correlated with alocally generated version of the pseudo-random noise code used by thetag. The cross correlator can determine the received phase of the signalto an accuracy of approximately one quarter of the chip period of the PNcode. The determined phase data output by the cross correlator 110 isthen passed directly to a data transmitter 114 which packages the datafor transmission to the position processor 4. However, with the phasemeasurement from the correlator, the position processor can onlydetermine the position of the tag to a resolution of approximately 10metres. Therefore, in this embodiment, the cross correlator 110 alsorecovers the carrier tones and outputs these to the DSP 112. In thisembodiment, the DSP processes these carrier tones in the same way as thetones were processed in the above embodiments. The phase informationextracted by the DSP 112 is then passed to the data transmitter 114 foronward transmission to the position processor 4 which operates in asimilar way to the position processor described above. The only maindifference is that the phase measurement obtained directly from thecross correlator 110 is used to provide a coarse position measurementand the phase measurements from the DSP 112 are used to provide anaccurate position measurement.

[0180] To distinguish between a plurality of mobile tags 2, the systemof the present embodiment allocates separate frequencies for each tag totransmit on. Alternatively, the PN code used in each tag could have beenmade different although this will complicate the structure of thereceivers as each will have to correlate the received signal with anumber of different locally generated PN codes. Provided that there issufficient coding gain, i.e. that the cross-correlation sum betweendifferent codes is low enough, the signals from each tag may overlap infrequency without causing interference. It is also possible to frequencyhop the system to provide additional robustness to narrow bandinterference, although spread spectrum systems have a very high inherentnoise rejection capability.

[0181] Modifications

[0182] Although it has been described above to use the tracking systemof the present invention to track horses in a horse race, the presentinvention is also applicable to dog racing, athletics, cycle racing andmotor racing for example. The tracking system would be most useful insports such as horse racing, athletics and dog racing as it is in theseraces that the small and unintrusive nature of the transmitter which theparticipant is required to wear or carry will be of greatest benefit.

[0183] In the above embodiments, three or four receivers were used totrack the position of one or more mobile tags. As those skilled in theart will appreciate, this number of receivers was used in order to beable to calculate the absolute two-dimensional position of the tagrelative to the receivers. However, if the position of the tag isconstrained then fewer receivers may be used. For example, two receiversmay be used in an embodiment similar to the first embodiment if the tagis constrained to move on one side of the receivers. Similarly, use ofany additional receivers can be used to provide a position measurementin three dimensions (i.e. in height as well as in the x and y horizontaldirections). As a further alternative, receivers may be deployed aroundthe side of the track and controlled in such a manner that only a few ofthe receivers nearest to the tag are used at any one time. In such anembodiment, a “handover” process to introduce and remove receivers fromthe tag position calculation could be used. Such a handover processcould take the form of an active system similar to those implemented incellular telephone networks or as a simple system of estimating from theposition and velocity which receivers will be closest and ignoring thedata received from the more distant receivers. Whilst the positiondetermining systems described above can operate with two or morereceivers, they preferably use as many receivers as possible in order toprovide redundancy in the position calculations.

[0184] In all of the embodiments described above, the receivers havebeen fixed and the transmitters have moved relative to the receivers. Asthose skilled in the art will appreciate, the receivers may also moveprovided their relative positions are known. However, such an embodimentis not preferred because of the complexity involved in maintainingknowledge of the positions of the different receivers. As a furtheralternative, the transmitter may be fixed and the receivers may moverelative to the transmitters. Such an embodiment could operate insubstantially the same way as the embodiment described above providedthe receivers are moved in unison.

[0185] As a further alternative, the function and operation of thetransmitters and the receivers may be reversed so that each tag becomesa receiver and each receiver becomes a transmitter. In this case, thesystem would operate in a similar manner to the system described in U.S.Pat. No. 5,045,861, except that each of the fixed transmitters wouldtransmit the multi-tone signal and each of the mobile tags would receivethe transmitted signal and either process the signals directly orforward the signals on to a remote position processor where the aboveprocessing techniques can be used to determine the relative position ofthe tag relative to the fixed transmitters. The way in which such anembodiment would operate will be apparent to those skilled in the artand a further description thereof shall be omitted.

[0186] In the above embodiments, each receiver received the signaltransmitted from each tag and calculated phase measurements which itthen passed to a central processing station. The central processingstation then calculated phase difference measurements and used thesephase difference measurements to calculate the position of the tagrelative to the receivers. In an alternative embodiment, the phasedifference calculations may be performed in the respective receivers.Such an embodiment is not preferred, however, since it increases theamount of processing that each receiver must perform.

[0187] In the above embodiments, each of the receivers received thesignal transmitted by each of the tags and processed the received signalto determine phase measurements for the signal. As those skilled in theart will appreciate, it is not essential for these phase measurements tobe carried out at the respective receivers. The processing may becarried out by the position processor or by some other intermediatecalculating station. All that the receivers have to do is provide a“snapshot” of the signal that they receive. The remaining processing canbe carried out elsewhere.

[0188] Where it has been described with reference to the specificembodiments to transmit the chirps within the frequency band from 2.4GHz to 2.485 GHz, this should not be viewed as limiting, anytransmission frequency can be used.

[0189] Although it has been described above with reference to the firstembodiment to provide a range of 300 metres and with respect to thefourth embodiment to provide a range of 3,000 metres, these rangesshould not be construed as limiting the present invention. The range ofthe tracking system may be determined by choosing appropriate frequencydifference pairs to cover the desired measurement area.

[0190] While it has been described above with reference to the aboveembodiment that the FPGA 10 will provide data describing the start phasefor each tone in a given chirp to the DDS 12, this is not essential. Asan alternative, the DDS 12 may be able to calculate the start phaseitself or to continue the generation of the signal and merely not tooutput it when it is not required and thus the start phase informationwould not be supplied by the FPGA 10 to the DDS 12.

[0191] In the above embodiments, the tones of each chirp weretransmitted alternately. As those skilled in the art will appreciate,this is not essential. The tones may be transmitted simultaneously or inany sequence. However, the alternate pulsing of the tones used in theabove embodiments allows simplification of the hardware in thetransmitter and the receivers because it is never necessary to deal withmore than one tone at any one time.

[0192] Although it has been described above with reference to the fourthembodiment to have a chirp structure as described with reference to FIG.15, other chirp structures and relative frequencies of tones within thechirp may be used. For example, a chirp having the following seven toneswith frequencies relative to a centre frequency of +0.1 MHz, 0 MHz, −10MHz, +2.5 MHz, −9.5 MHz, 0 MHz and +0.5 MHz respectively may be used.Such a chirp structure would provide one measurement with a differenceof 0.1 MHz, two measurements with a difference of 0.5 MHz, fourmeasurements with a difference of 2.5 MHz and eight measurements with adifference of 10 MHz. This chirp structure provides a more gradualtransition between different wavelengths than the chirp structuredescribed with reference to the fourth embodiment. It also comprisesmore tones and a longer actual measurement period (assuming that thelength of each tone within the chirp is unchanged) thereby providing asmall improvement in signal to noise ratio (SNR). The use of 8difference frequencies of around 10 MHz would provide an improvedresolution over the fourth embodiment. Double the number of frequencydifferences and half the wavelength should yield a four timesimprovement. The use of two intermediate stages at 0.5 MHz and 2.5 MHzshould improve the robustness of position acquisition by increasingposition resolution thus reducing cycle count ambiguities. Also, the useof the 2.5 MHz difference will provide an intermediate fallback in theevent that signal quality is too poor to use the 10 MHz differencesignals.

[0193] Although it has been described above with reference to the fourthembodiment that each chirp contains a tone at a frequency f_(ID) whichfrequency is unique for each tag, it would alternatively be possible toincorporate a data carrying tone into the chirp onto which dataincluding the tag ID could be modulated using a conventional datamodulation technique. This arrangement would also provide for additionaldata to be transmitted with each chirp, for example battery power orother operating conditions.

[0194] Although it has been described above with reference to thespecific embodiments to use a duration of 0.3 ms for each positioningtone within a chirp, this is not restrictive and other durations arepossible. Increasing the tone duration provides an increase in SNR forthat tone, and if all tones are extended in duration such that the chirphas greater duration, then the chirp SNR will also be increased.Decreasing the tone duration provides a smaller collision probabilitywhen multiple tags are transmitting on the same frequencies as if eachchirp is shorter in duration, then there is less chance of it collidingin time with another chirp.

[0195] Although it has been described above in the specific embodimentsthat the DDS generating the tones in the tags performs a simpleswitching operation between the tones, a more complex switchingoperation could be utilised. For example, where a “hard” ON-OFF switchis used between the tones this has the effect of broadening the spectrumas though it were an FSK system with extended side lobes onapproximately 30 kHz spacing (assuming a 0.3 microsecond switchingrate). In the case of a system where only a small number of tags aretransmitting this is not a serious problem, however in a system where alarge number of tags are transmitting this could cause significantinterference between the chirps from different tags. It is thereforepossible to shape each tone transmission. For example, Gaussian shapingmay be used such that the amplitude of each tone will be Gaussian shapedacross the duration of the tone. If a Gaussian filter with a bandwidthequivalent to one tone period with a Gaussian shaping factor of between0.7 and 1.0 is used, the central 0.2 milliseconds of the 0.3milliseconds tone has sufficient amplitude to be utilised for phasemeasurement, whilst achieving good suppression of spectral side lobescaused by tone switching. Such chirp shaping would not improve systemaccuracy significantly, however it would enable the use of a largenumber of tags. The shaping would also help to ensure that thetransmitted spectrum is contained within the desired band therebyproviding some SNR improvement. It would also help with compliance withIEEE 802.11 Regulations and reduce the likelihood of intermodulationdistortion in the transmitter.

[0196] Where it has been described above with reference to the fourthembodiment to pass the sampled, mixed and decimated input data directlyto a complex FFT, further processing (such as windowing the data) couldbe introduced to improve performance. For example, without windowing thedata or overlapping the data the discrimination between FFT bins is nothigh and the system is less well suited for a situation where chirpsfrom different tags are being received simultaneously in differentfrequency bins and good near-far performance needs to be attained. Forexample, it would be possible to implement a 14 MHz system referenceclock, and a 14 bit ADC which gives up to 80 dB dynamic range, allowingfor 54 dB of near-far separation allowing for a minimum of 26 dB SNR.This provides a 500:1 physical range under unobstructed line of sightconditions which, for example, could be implemented as a 6 metre to3,000 metre range capability. This would produce a data stream into thesignal processing block at 28 megasamples per second in I and Q having abandwidth of 22.4 MHz. A 1024 point FFT operating on that data streamwould give bin widths of approximately 27 kHz, since the chirp tones arearranged to rely on approximately 50 kHz spacing which is equal to twicethe bin width it would be necessary to achieve a bin+1 attenuation ofclose to 80 dB. Even with careful data windowing this is difficult toachieve using an FFT. An alternative to the FFT is to use a directimplementation such as an FIR filter bank.

[0197] Although it has been described above with reference to thespecific embodiments to use a fixed tag 5 located at a known locationwithin the domain of the tracking system, it is possible to modify thelocation of the fixed tags such that each fixed tag is a single jointentity with a corresponding one of the receivers. This results in thepositions of the fixed tags and the receivers becoming the same in thelocation mathematics but is also means that no separate fixed tags arerequired when deploying the system. This also means that there are asmany fixed tags as there are receivers. However, with the fixed tags andthe receivers each sharing the same clock, the calibration of thereceiver network to a single network clock (that can be one, any or someaverage of the receiver clocks) is much simpler than the techniquedescribed above with reference to the second embodiment. This,therefore, reduces the computational load on the position processorsince sets of position equations are now only received for each receiverand each measurement frequency and it is not necessary to multiply bythe number of fixed tags. In addition, the use of a whole network offixed tags gives a mesh of propagation paths cris-crossing the area ofcoverage with complete inter-linkage between receivers. Using the entiremesh of paths to obtain the network reference clock will improve systemperformance.

[0198] Although it has been described above with reference to thespecific embodiments that the link between the receivers and theposition processor is established using a wireless TCP/IP network, thatcase is not limiting and any suitable cabled or wireless network usingany suitable network protocol may be used to establish the receiver toposition processor links.

[0199] Although it has been described above with reference to the firstto fifth embodiments to use an FPGA and DDS, and with reference to thesixth embodiment to use a signal generator, to generate the tone signalswithin the tags, any method of generating the tones in a stablephase-continuous manner may be used. An example of an alternative methodwould be to use a crystal oscillator at each of the required frequenciesto generate the tones.

[0200] In the above embodiments, the or each transmitter transmitted amulti-tone signal in which the frequency spacing between the tones wasknown in advance. As those skilled in the art will appreciate, this isnot essential. The results of the FFT analysis performed in thereceivers can identify the frequencies of the transmitted tones andhence identify the spacing therebetween. Similarly, in the frequencyhopping embodiment described above, the frequency hopping schedule doesnot need to be known in advance and can be determined directly from theFFT results from each of the receivers.

[0201] Although the embodiments described above have used computerapparatus and processes performed in computer apparatus, the inventionalso extends to computer programs, particularly computer programs on orin a carrier, adapted for putting the invention into practice. Theprogram may be in the form of source code, object code, a codeintermediate source and object code such as any partially-compiled form,or in any other form suitable for use in the implementation of theprocesses according to the invention. The carrier may be any entity ordevice capable of carrying the program. For example, the carrier maycomprise a storage medium, such as a ROM, for example a CD-ROM or asemi-conductor ROM, or a magnetic recording medium, for example a floppydisc or hard disc. Further, the carrier may be a transmissible carriersuch as an electrical or optical signal which may be conveyed viaelectrical or optical cable or by radio or other means.

1. A position determining system comprising: a transmitter beingoperable to transmit a signal comprising first and second frequencycomponents having a frequency spacing therebetween; a plurality ofreceivers having known relative positions, each being operable toreceive the signal transmitted from the transmitter; means forprocessing the signal received at each receiver to determine, for eachreceived signal, a phase measurement for the first frequency componentand a phase measurement for the second frequency component; means forcalculating a phase difference measurement for each received signal fromthe determined phase measurements for the corresponding received signal;and means for determining the relative position between the transmitterand the receivers on the basis of the calculated phase differencemeasurements for the received signals and the known relative positionsof the receivers.
 2. The system of claim 1, wherein a separateprocessing means is provided for each receiver which is located at thecorresponding receiver and which is operable to determine the phasemeasurements for the signal received at the corresponding receiver. 3.The system of claim 1 or 2, wherein said calculating means and saiddetermining means are located within a central processing station, andwherein said processing means is operable to transmit said phasemeasurements to said central processing station.
 4. The system of anypreceding claim, wherein the transmitter is operable to transmit thefirst frequency component and the second frequency componentalternately.
 5. The system of any preceding claim, wherein thetransmitter is operable to transmit pulses of said signal, wherein saidprocessing means is operable to determine a phase measurement for thefirst and second frequency components during each pulse of saidtransmitted signal, wherein said calculating means is operable tocalculate a phase difference measurement for each received signal duringeach pulse and wherein said determining means is operable to determine aposition of the transmitter at the time of each pulse on the basis ofthe calculated phase difference measurements for the received signalsfor the corresponding pulse and the known relative positions of thereceivers.
 6. The system of claim 5, wherein the transmitter is operableto maintain phase continuity between transmitted pulses of said signal.7. The system of claim 5 or 6, wherein the transmitter comprises asingle clock from which said first and second frequency components arederived.
 8. The system of claim 5, 6 or 7, wherein said processing meansis operable to determine the phase of each frequency component at eachof a plurality of different times during each pulse and wherein saiddetermined phase measurement for each frequency component comprises aphase offset value corresponding to the phase of the respectivecomponent at one of said times and a phase slope measurement indicativeof the rate at which the determined phase of said frequency componentchanges during each pulse.
 9. The system of claim 8, wherein saidprocessing means is operable to perform repeated frequency analysis ofsaid received signals to determine said phase measurements.
 10. Thesystem of claim 9, wherein said processing means is operable to performrepeated frequency transforms of the received signals to determine saidphase offset measurement and said phase slope measurement.
 11. Thesystem of claim 9 or 10, wherein said processing means repeatedlyperforms said frequency analysis on each received signal and furthercomprises means for storing a reference pattern representative of anexpected result of the frequency analysis of the signal transmitted bythe transmitter; and means for comparing the results of said repeatedfrequency analysis with said reference pattern to identify a receivedpulse of the transmitted signal.
 12. The system of any of claims 8 to11, wherein said calculating means is operable to calculate a phasedifference measurement for both said phase offset measurement and saidphase slope measurement.
 13. The system of any of claims 8 to 12,further comprising means for determining the posit-on of saidtransmitter between or after transmitted pulses by interpolating orextrapolating from determined positions at the pulses.
 14. The system ofany preceding claim, comprising a plurality of transmitters, eachoperable to transmit a respective signal comprising first and secondfrequency components, wherein said plurality of receivers are operableto receive the signal transmitted from each transmitter, wherein saidprocessing means is operable to process the signal received at eachreceiver from each transmitter to determine, for each received signal,said phase measurement for the first frequency component and said phasemeasurement for the second frequency component, wherein said calculatingmeans is operable to calculate a phase difference measurement for eachreceived signal and wherein said determining means is operable todetermine the position of each transmitter on the basis of thecalculated phase difference measurements for the received signals fromthe corresponding receiver and the known relative positions of saidreceivers.
 15. The system of claim 14, wherein each transmitter isoperable to transmit on different frequencies.
 16. The system of claim14 or claim 15, wherein each transmitter is operable to transmit pulsesof said signal at a different repeat interval.
 17. The system of anypreceding claim, wherein the or each transmitter is operable to transmitsaid signal comprising said first and second frequency components havinga predetermined frequency spacing therebetween.
 18. The system of anypreceding claim, wherein the or each transmitter comprises means forchanging the transmission frequency of said first and second frequencycomponents.
 19. The system of claim 18, wherein said changing means isoperable to maintain the same frequency spacing between said first andsecond frequency components.
 20. The system of claim 18 or 19, whereinthe changing means is operable to change the transmit frequenciesaccording to a predetermined schedule.
 21. The system of any precedingclaim, wherein the transmitter is operable to transmit said signal as aspread spectrum signal.
 22. The system of claim 21, wherein thetransmitter is operable to generate said spread spectrum signal bycombining said signal with a pseudo-noise code and wherein saidprocessing means comprises a correlator for correlating the receivedsignal with a copy of the pseudo-noise code to determine said phasemeasurement for each of said first and second frequency components. 23.The system of claim 21 or 22 when dependent on claim 14, wherein eachtransmitter uses a pseudo-noise code unique to that transmitter.
 24. Thesystem of any preceding claim, wherein the transmitter is operable totransmit a signal comprising first, second and third frequencycomponents, each having a frequency spacing from the other frequencycomponents, wherein said processing means is operable to determine aphase measurement for each frequency component, wherein said calculatingmeans is operable to determine a plurality of phase differencemeasurements for each received signal from the determined phasemeasurements for the first, second and third frequency components of thereceived signal, and wherein said determining means is operable todetermine the relative position of the transmitter on the basis of thecalculated phase difference measurements for the received signals andthe known relative positions of the receivers.
 25. The system accordingto claim 24, wherein the frequency spacing between the first and secondfrequency components is greater than the frequency spacing between thesecond and third frequency components and wherein the phase differencemeasurements obtained from the phase difference measurements of thefirst and second frequency components are operable to provide a coarseposition measurement and wherein the phase difference measurementsobtained from the phase measurements of the second and third frequencycomponents are used to determine a fine position measurement.
 26. Thesystem of claim 25, which is operable to determine the relative positionof said transmitter over a predetermined range and wherein saidfrequency spacing between said first and second frequency components ischosen so that said coarse position measurement provides an absoluteposition measurement within said range.
 27. The system of claim 26,wherein the frequency spacing between said second and third frequencycomponents is determined so that said fine position measurement includesa cyclic ambiguity within said range and wherein said coarse positionmeasurement is used to resolve said cyclic ambiguity.
 28. The system ofclaim 25, 26 or 27, wherein said determining means is operable todetermine the position of the transmitter using an iterative numericaltechnique, with the coarse position measurement being used to initialisethe iterative processing to determine said fine position measurement.29. The system of any preceding claim, wherein said receivers areunsynchronised and further comprising a reference transmitter whoseposition relative to said receivers is known and operable to transmit areference signal having first and second frequency components with afrequency spacing therebetween, wherein said plurality of receivers areoperable to receive the reference signal transmitted from the referencetransmitter, wherein said processing means is operable to process thereference signal received at each receiver to determine for eachreceived reference signal, a phase measurement for the first frequencycomponent and a phase measurement for the second frequency component,wherein said calculating means is operable to calculate a phasedifference measurement for each received reference signal from thedetermined phase measurements for the corresponding received referencesignal and further comprising: means for determining a respectivecalibration value for each receiver from the calculated phase differencemeasurements for the received reference signals, the known relativepositions of the receivers and the known relative position of thereference transmitter; and means for adjusting said phase measurementsusing said calibration values to account for the lack of synchronisationof said receivers.
 30. The system of claim 29, wherein said adjustingmeans is operable to adjust said phase difference measurements usingsaid calibration values.
 31. The system of claim 29 or 30, comprising aplurality of said reference transmitters.
 32. The system of claim 31,wherein each reference transmitter is located at a correspondingreceiver.
 33. The system of any preceding claim, wherein saidtransmitter is a transmit-only transmitter and operates asynchronouslywith respect to said receivers.
 34. The system of any preceding claim,further comprising a plurality of tracking loops for tracking andsmoothing each of the calculated phase difference measurements.
 35. Thesystem of claim 34, wherein each tracking loop comprises a phase lockedloop.
 36. The system of any preceding claim, wherein said determiningmeans is operable to determine a two-dimensional position of saidtransmitter.
 37. The system of claim 36, wherein three receivers areprovided and wherein said determining means is operable to determine theabsolute position of said transmitter in two dimensions.
 38. The systemof any of claims 1 to 36, wherein said determining means is operable todetermine the position of said transmitter in three dimensions.
 39. Thesystem of any preceding claim, wherein said determining means isoperable to determine the relative position between the transmitter andthe receivers on the basis of the distance between the transmitter andeach receiver from said phase difference measurements.
 40. The system ofany preceding claim, comprising a reference transmitter whose positionis known relative to said receivers and operable to transmit a signalcomprising first and second frequency components having a frequencyspacing therebetween; and wherein said receivers, said processing meansand said calculating means are operable to process a signal from saidreference transmitter to generate calibration values for use incalibrating the phase measurements from said transmitter.
 41. The systemof claim 40, wherein said calibration values are repeatedly updated andused to dynamically alter the phase measurements from the transmitter inorder to reference the measurements from the transmitter to a clockwithin said reference transmitter.
 42. A transmitter for use in thesystem according to any preceding claim, comprising: a clock forgenerating a clock signal; means for receiving the clock signal and forgenerating therefrom a plurality of frequency components having afrequency spacing therebetween; and means for transmitting a signalcomprising said plurality of frequency components.
 43. The transmitterof claim 42, wherein said generating means is operable to generatepulses of said plurality of frequency components whilst maintainingphase continuity between the pulses.
 44. The transmitter of claim 42 or43, wherein said generating means is operable to generate said pluralityof frequency components in sequence.
 45. The transmitter of any ofclaims 42 to 44, wherein said generating means comprises a frequencysynthesiser which is operable to generate frequencies within apredetermined frequency band and a programmable memory device whichstores data defining the frequencies to be synthesised by saidsynthesiser.
 46. The transmitter of claim 45, wherein said programmablememory device comprises data defining the start time and stop time ofeach frequency component synthesised by said synthesiser.
 47. Thetransmitter of claim 45 or 46, wherein said programmable memory devicecomprises a field programmable gate array.
 48. The transmitter of any ofclaims 45 to 47, wherein said synthesiser is a digital synthesiser andfurther comprising a digital to analogue converter for convertingdigital samples output from said digital synthesiser to generate acorresponding analogue frequency component.
 49. A position processor fordetermining the position of a transmitter relative to a plurality ofreceivers, the receivers having known relative positions and beingoperable to receive a signal comprising first and second frequencycomponents having a frequency spacing therebetween transmitted from thetransmitter, the apparatus comprising: means for receiving a pluralityof sets of phase measurements, each set associated with a respective oneof the receivers and each set comprising a phase measurement for thefirst frequency component and a phase measurement for the secondfrequency component of the signal received at the correspondingreceiver; means for calculating a phase difference measurement for eachset of phase measurements; and means for determining the relativeposition of the transmitter on the basis of the calculated phasedifference measurements and the known relative positions of thereceivers.
 50. A position determining method for determining therelative position between a transmitter and a plurality of receivers,with the relative position of the receivers being known, the methodcomprising the steps of: transmitting from the transmitter a signalcomprising first and second frequency components having a frequencyspacing therebetween; receiving at each receiver the signal transmittedby the transmitter; processing the signal received at each receiver todetermine, for each received signal, a phase measurement for the firstfrequency component and a phase measurement for the second frequencycomponent; calculating a phase difference measurement for each receivedsignal from the determined phase measurements for the correspondingreceived signal; and determining the relative position between thetransmitter and the receivers on the basis of the calculated phasedifference measurements for the received signals and the known relativepositions of the receivers.
 51. A position determining systemcomprising: a tag and a plurality of base stations, the tag beingmovable relative to the base stations and the position of each basestation relative to the other base stations is known; wherein the tagand the plurality of base stations are arranged so that upon thetransmission of a signal comprising first and second frequencycomponents having a frequency spacing therebetween from the tag to thebase stations or from the base stations to the tag, there is generated aplurality of received signals each associated with a respectivetransmission path between a respective base station and the tag; meansfor processing each received signal to determine a corresponding phasemeasurement for the first frequency component and a corresponding phasemeasurement for the second frequency component; means for calculating aphase difference measurement for each received signal from thecorresponding determined phase measurements; and means for determiningthe relative position of the tag and the base stations on the basis ofthe calculated phase difference measurements for the received signalsand the known relative positions of the base stations.
 52. The system ofclaim 51, wherein said tag transmits said signal and wherein each ofsaid base stations receives said signal.
 53. The system of claim 51,wherein each of said base stations transmits said signal and whereinsaid tag receives each signal.
 54. The system according to claim 53,wherein said processing means, calculating means and determining meansare provided in the tag.
 55. The system of claim 53, wherein saiddetermining means is provided in a central processing system and whereinsaid tag is operable to transmit signals to said central processingsystem, which signals depend upon the signals received from said basestations.
 56. The system of claim 55, wherein said processing means andsaid calculating means are provided within said central processingsystem.
 57. Processor-implementable instructions for programming aprogrammable computer device to become configured as the positionprocessor of claim
 49. 58. Processor-implementable instructions forcausing a programmable computer device to become configured as atransmitter according to any of claims 42 to
 48. 59. A receiver forreceiving a signal comprising first and second frequency componentstransmitted by a tag, the receiver comprising: means for receiving thesignal transmitted by the tag; means for performing a repeated frequencyanalysis of the received signal to obtain a plurality of phasemeasurements for each frequency component in the received signal; andmeans for processing the plurality of phase measurements for eachreceived frequency component to determine a phase measurement for eachtone; and means for outputting said phase measurements for transmissionto a central position processor.
 60. The receiver of claim 59, whereinsaid processing means is operable to determine a phase offsetmeasurement and a phase slope measurement for each frequency component.61. Processor implementable instructions for causing a programmableprocessor device to become configured as the receiver of claim 59 or 60.